Device for switching a semiconductor-based switch and sensor for detecting a current change velocity at a semiconductor-based switch

ABSTRACT

An electric circuit includes a semiconductor-based switch with a first terminal and a second terminal for conducting a power current. Further, the electric circuit includes a sensor for detecting a current change velocity of the power current flowing through the semiconductor-based switch. The sensor includes an insulating foil configured to be connected to the first or second terminal of the semiconductor-based switch and an inductance arranged on the insulating foil on a side of the same that is arranged opposite to a side facing the semiconductor-based switch during a measurement operation of the sensor in order to detect a magnetic field generated by the power current to provide a measurement voltage based on the detected magnetic field. The inductance is spaced apart from the semiconductor-based switch at least by the insulating foil, such that contactless measurement is enabled. The insulating foil includes a mounting portion.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of copending InternationalApplication No. PCT/EP2015/058185, filed Apr. 15, 2015, which isincorporated herein by reference in its entirety, and additionallyclaims priority from European Application No. EP 14165895.5, filed Apr.24, 2014, and German Application No. 102014214246.6, filed Jul. 22,2014, which are all incorporated herein by reference in their entirety.

The present invention relates to devices for switching asemiconductor-based switch and to a sensor for detecting a currentchange velocity of a power current flowing through a semiconductor-basedswitch.

BACKGROUND OF THE INVENTION

Nowadays, so-called power converters are used for converting electricenergy. The same allow the connection of energy sources with givenvoltages, currents and frequencies to energy sinks or loads. Here, theconverter converts electric energy of one form, such as alternatingcurrent (AC), direct current (DC) or a mixed form thereof into adifferent form, in a single or multiple stages. Generally, the converterconsists of semiconductor-based switches connected in the necessitatedtopology, for example as H bridge inverter or as 3-phase 6-pulseinverter. The most frequently used electronic switches are insulatedgate bipolar transistors (IGBT) or metal oxide semiconductor fieldeffect transistors (MOSFET). These semiconductor switches are controlledvia so-called gate drivers and hence represent the interface between thepower switch and the small-signal main control of the converter.

The requirements for such a gate driving unit (GDU) are the following:small installation space (usually determined by the form factor of thepower switch package), sufficient driving power (when activating ordeactivating the switches, extreme pulse energies are necessitated forreliable switching), ensuring small switching losses of the powerswitches (switching losses are frequently the main focus when optimizingconverters), ensuring small conduction losses of the power switches,little effort for driver adaptations when using new or different powerswitches (plug and play), flexible scalable GDU for differentapplications (reusability, avoiding design variations), short death ordelay time in the signal path of the converter control, highconfigurable degree of protection and/or interference radiationconforming to standards in any operating point if possible.

FIG. 10 illustrates the standard topology of a GDU with its functionalblocks. Switching on a semiconductor-based switch 12 is performed bymeans of a voltage source V_(on) and a transmission factor k1. Thereby,a gate current I_(Gon) is induced. Switching off is performed with avoltage sink V_(off) and the transmission factor k2. Thereby, the gatecurrent I_(Goff) is induced. In the error case “overcurrent”, analternative switch-off path is used via the transmission factor k2 bywhich a gate current I_(Goff,err) is induced. This enables smallinstallation space, ensuring low conduction losses in the power switch12, inexpensive production and a basic degree of protection.

It is a disadvantage of such an implementation that the driving power isnot optimal for each switching time/operating time, that the switchinglosses can only be optimized for one operating point (voltage, loadcurrent, temperature). Further, a tradeoff has to be found between shortdeath/delay time and di/dt behavior, du/dt behavior as well as powerlosses in the tail phase. Further, it is disadvantageous that theswitching sequence does not compensate any component leakages, that higheffort is necessitated when using new or different power switches forchanging the topology or for adapting the switching behavior bycomponent changes, that no flexible adaptation or scalability is givenas well as no high or no configurable degree of protection is possible.Interference radiation conforming to standard heavily influences thelosses and is not suitable for a partial load at the switch 12.

The topology in FIG. 10 is also described as single-stage resistivelycontrolled or as GDU resistive with SoftTurnOff.

FIG. 11 shows a basic structure of a single-stage/multi-stageanalog-controlled GDU with active change of the gate dropping resistor.Here, two and up to, for example, a maximum of eight switching stagesare useful for switching on as well as switching off thesemiconductor-based switch 12. Mostly, two or three stages are known.Switching on is performed with one voltage source V_(on) each and thetransmission factors k1,1 . . . k,n. Thereby, the gate current I_(Gon,x)is induced in different paths with x=1, . . . ,n. Controlling thesequence can be performed directly by the superordinate controlinstance. Alternatively, the control can be performed with a configuredtime control on the GDU.

Switching off is performed with a voltage sink V_(Off) and thetransmission factors k2,1 . . . k2,n. Thereby, the gate currentI_(Goff,x) is induced with x=1, . . . ,n. The sequence can be performeddirectly by the superordinate control instance. Alternatively, thesequence can be performed with a configured time control on the GDU. Inthe error case “overcurrent”, an alternative switch-off path via thetransmission factor k3 is used. This offers the advantage of improvedadaptation of the driving power for each switching time, improvedoptimization between short death/delay time, di/dt behavior, du/dtbehavior and power losses in the tail phase, reduced impact on the powerlosses with interference radiation conforming to standards, ensuringsmall conduction losses in the power switch as well as a basic degree ofprotection. In other words, FIG. 11 shows a GDU having variable gateresistors.

Disadvantages of such an embodiment are an increased need forcomponents, a more complex sequence control as well as additional(galvanically isolated) control channels. This results in increasedcosts. Further, it is disadvantageous that switching losses can beoptimized only for one operating point (voltage, load current,temperature), that the switching sequence does not compensate anycomponent leakages, that high effort results when using new/differentpower switches, that no flexible adaptation/scalability of the circuitis possible, that no high or even configurable degree of protection canbe obtained. Due to lack of feedback, switching sequence drift ispossible.

FIG. 12 shows a basic structure of a GDU having two voltage levels.Generally, this can be referred to as GDU having multiple voltagelevels. Two voltage levels each are documented for switching on andswitching off the semiconductor-based switch 12. Switching on isperformed starting with a voltage source V_(On2) and reduced gatecurrent I_(Gon). Then, switching to voltage source V_(On1) and maximumgate current I_(Gon) follows. Controlling the sequence can be performeddirectly by the superordinate control instance. Alternatively, thesequence can be performed with a configured time control on the GDU.Switching off takes place starting with a voltage sink V_(Off2) andreduced gate current I_(Goff). Then, switching to the voltage sinkV_(Off1) and maximum gate current I_(Goff) follows. The sequence can beperformed directly by the superordinate control instance. Alternatively,the sequence can be performed with a configured time control on the GDU.In the error case “overcurrent”, an alternative switch-off path via thetransmission factor k3 is used.

This provides the advantage of improved adaptation of the driving powerfor each switching time, improved optimization between short death/delaytime, di/dt behavior, du/dt behavior and power losses in the tail phase.Further, interference radiation conforming to standards can be obtained,which influences the switching losses to a lesser degree. Lowerconduction losses in the power switch can be ensured and a basic degreeof protection can be obtained.

It is a disadvantage that increased need for components, more complexsequence control as well as additional (galvanically isolated) controlchannels result which result in increased costs. Switching losses canonly be optimized for one operating point (voltage, load current,temperature). Further, the switching sequence does not compensate anycomponent leakages. A high effort results when using new/different powerswitches, no flexible adaptation/scalability of the system is possible,and no high or even configurable degree of protection can be obtained.Switching sequence drift due to lack of feedback can occur. Implementinga second voltage level causes additional losses, depending on therealization.

FIG. 13 shows the basic structure of a single-stage/multi-stageanalog-controlled GDU. Different documented extension levels exist forthis principle. However, all of them are based on returning thecollector potential/drain potential for detecting the duc/dt and dud/dt,respectively (derivative drain voltage over time) as well the voltage ofthe module-internal leakage inductance between auxiliary emitter/sourceand load emitter/source as a measure for dic/dt (derivative collectorcurrent over time) and did/dt, respectively (derivative drain currentover time). Switching on is generally performed via a voltage sourceV_(On) and the base resistor (impedance) k1. Thereby, the gate currentI_(Gon) is induced. Depending on the implementation, in the switchingedge, reduction of the gate current follows during the di/dt phase and arise/reduction during the du/dt phase. Switching off is generallyperformed by a voltage source V_(Off) and the base impedance k2.Thereby, the gate current I_(Goff) is induced. Depending on theembodiment, in the switching edge, reduction of the gate current followsduring the di/dt phase and a rise/reduction during the du/dt phase. Inthe error case “overcurrent”, an alternative switch-off path via theimpedance k3 is used.

This enables better adaptation of the driving power for each switchingpoint/operating point and good optimization between short death/delaytime, di/dt behavior, du/dt behavior and power losses in the tail phase.Interference radiation conforming to standards influences switchinglosses to a lesser extent. Further, low conduction losses in the powerconductor can be ensured. The concept offers a basic degree ofprotection and good adaptation to components (i.e., power switches) andedge parameter changes by feedback signals.

The disadvantages of this concept are that a significantly increasedneed for components results, mostly expensive analog components having ahigh bandwidth are necessitated, increased losses occur in the GDU bycomponents having a high bandwidth, that feedback circuits causeadditional losses due to intervention, that increased costs occur, thatswitching losses can only be optimized for one operating point (voltage,load current, temperature), that the switching sequence does notcompensate any component leakages, that high effort results when usingnew/different power switches, that no flexible adaptation or scalabilityof the concept is possible, that merely a configurable but no highdegree of protection can be obtained, that a high voltage potential(collector) necessitates installation space on a driver area, since airand creeping distances have to be considered, that high voltagecomponents have great leakages, that no adaptation of the control planeis possible by interventions of the feedbacks, that the usage of themeasured parasitic leakage inductance of the power switch is noprecisely specified characteristic of the power switch itself, that withsmall leakage inductances no direct processing of the signal is possibleand that the direct intervention with feedback comprises a phase shiftsuch that the intervention, depending on the impedances in the gateline, only has a limited or no effect in fast switching processes andcan result in undesirable oscillations.

FIG. 14 shows an embodiment of a single-stage/multi-stage digitallycontrolled GDU. The same serves to synchronize parallel power switches.Wth the help of a di/dt sensor, the beginning of the current rise(current change) during switching on as well as the beginning of thecurrent drop during switching off is detected. Differences betweenswitching on start and switching off start are stored and processed withfurther data in a DSP instance (DSP=digital signal processor). Thereby,delays for the respective switch-on time and switch-off time aredetermined for the next switching time and transmitted to the FPGA(FPGA=Field Programmable Gate Array).

This concept offers, among others, the advantages that low conductionlosses in the power switch are ensured, that a basic degree ofprotection can be obtained, that a good adaption to the component andedge parameter changes by feedback signals are enabled, that goodsymmetry of the switching edges is enabled, that no additional lossesoccur by direct intervention of the feedback circuits in the gatecurrent, that good adaptation of the switching time to componentleakages of the power switches is enabled and that module-independentdetection of the current change velocity di/dt is possible.

The disadvantages of this concept are a significantly increased need forcomponents due to two programmable instances, that mostly expensiveanalog components having a high bandwidth are necessitated, thatincreased losses occur in the GDU due to two programmable instances,that increased costs can result when using high-performance DSP and/orFPGA, that no optimization of the actual switching edge curve isnecessitatedpreferential direction/possible, that switching losses canonly be optimized for one operating point (voltage, load current,temperature), that the switching sequence does not compensate anycomponent leakages, that a high effort results when using new/differentpower switches, that no flexible adaptation/scalability is possible andthat merely a configurable but no high degree of protection can beobtained.

FIG. 15 shows a further embodiment of a digital control. In thisembodiment stage, up to n=7 purely resistive switch-on paths as well asup to n=7 purely resistive switch-off paths are implemented. Via aconfiguration, a desired operational sequence is programmed into theFPGA. Additionally, further operating parameters, such as intermediatecircuit voltage, maximum collector current, switching frequency, IGBTtype and power partial topology are stored. They all serve to configurethe protective functions. The protective functions include two-stagedi/dt detection and four-stage voltage collector emitter VCE detection.Switching on is performed by means of a time controlled state machineactivating the configured resistive switching-on paths one after theother. The VCE and di/dt states are merely processed for short circuitand desaturation monitoring.

Switching off is also performed in a time-controlled manner via timingvia the configured resistive switch-off paths.

Advantages of the concept are ensuring low conduction losses in thepower switch 12, a high configurable degree of protection, a lowhardware change effort for driver adaptations when using new/differentpower switches in terms of plug and play, a flexible scalable GDU fordifferent applications as well as optimization of the actual switchingedge curve for (but only one) operating point (voltage, load current,temperature).

However, it is a disadvantage of this concept that a significantlyincreased need for components results by a total of 14 switching paths,wherein also significantly increased costs result by 14 switching paths.The concept has poor efficiency as never all paths are used at any time.Further, great space requirements result due to a plurality of controlpaths and feedback paths. The usage of the parasitic leakage inductanceof the power switch is no specifically specified characteristic of thepower switch resulting in measurement inaccuracies. The conceptnecessitates signal processing of the leakage inductance due to the lowsignal amplitudes. The usage of high-performance DSP and FPGA results inincreased costs when implementing the concept. Thus, the switchinglosses can merely be optimized for one operating point (voltage, loadcurrent, temperature). Further, the switching sequence does notcompensate any component leakage.

There are further digital approaches, which essentially allow aconfiguration of different resistive switch-on and switch-off paths,also during runtime.

A further approach includes a completely closed control loop withdigital control core. Detecting the data is performed with fast analogcomponents and subsequent digitalization. The digitized data areprocessed in the computer core, and again analogized by means of adigital-to-analog converter (DAC) in order to be passed on to the outputstages as control information. Here, the main problem is the delay dueto the signal processing times. Thus, directly intervening into thedynamic switching behavior is only possible with very slow edge curves.

In summary, the main problem of all controlled methods, independent ofwhether the same are analog or digital is the lack of adaptation to theoperating point of the power unit. Optimization can only take place forone or a few operating points. Some component leakages and parametervariances of the power switches cannot be compensated.

All analog-controlled methods have the problem, as long as they havedirect influence on the current switching edge, that they necessitatevery expensive energy-hungry components having a high bandwidthresulting in a low energy efficiency. The intervention is performed byinverse feedback into the gate output stage. Thus, additional controllosses are generated.

In the di/dt control by means of leakage inductance between auxiliaryemitter/source and load emitter/source, great deviations exist of thisparameter of the power module that is not specified in detail and/or notguaranteed. Additionally, a ground loop is interspersed. Since theleakage inductance is generally very small and continuously optimized,no direct processing can be performed.

Returning and processing the auxiliary collector/drain potential forprotective functions and/or a du/dt control necessitates usage ofcomponents having high electric strength, frequently even in a cascadedmanner. The same are mostly expensive and frequently too inaccurate forsignal processing since the same are not operated in the optimumoperating point. Apart from that, looping-in this potential results in alarge unused installation area due to the necessitated voltageclearances.

Further, the approach of analog control loops is only developed andqualified for one topology/application for exactly one power switch.Adapting the topology to other or second source switches (second sourceis, for example, a switch of another producer identical in design) withslightly amended characteristics necessitates a hardware variation ofthe topology and hence a new design of the topology.

The problem of known digitally controlled solutions is also the lack ofindependent adaptation to the current operating point or the componentleakage. Merely several resistive paths can be selected and used. Unusedpaths represent unused areas and still cause component costs since thesame are populated.

SUMMARY

According to an embodiment, an electric circuit may have: asemiconductor-based switch with a first terminal and a second terminalfor conducting a power current; a sensor for detecting a current changevelocity of the power current flowing through the semiconductor-basedswitch, the sensor having: an insulating foil configured to be connectedto the first or second terminal of the semiconductor-based switch; andan inductance arranged on the insulating foil on a side of the same thatis arranged opposite to a side facing the semiconductor-based switchduring a measurement operation of the sensor in order to detect amagnetic field generated by the power current and to provide ameasurement voltage based on the detected magnetic field; wherein theinductance is spaced apart from the semiconductor-based switch duringthe measurement operation at least by the insulating foil, such thatcontactless measurement of the current change velocity by the sensor isenabled; wherein the insulating foil further includes a mountingportion; wherein the first or second terminal of the semiconductor-basedswitch is connected to the mounting portion of the insulating foil;wherein the first or second terminal of the semiconductor-based switchis connected to a current conductor, wherein the mounting portion atleast partly surrounds the current conductor; and wherein the first orsecond terminal is connected to the current conductor by means of ascrew connection, wherein the mounting portion of the sensor is a screwhole, and wherein the sensor is arranged between a first and a secondelement of the screw connection and is fixed by means of the screwconnection.

According to another embodiment, a device for switching asemiconductor-based switch, may have: an inventive electric circuit; acontrollable voltage source that is configured to provide a time-varyingvoltage potential; a controllable resistive circuit having at least twoohmic resistances connected in parallel that are controllable such thatat least three resistance values of the parallel connection result; acontrol device that is configured to control the controllable voltagesource and the controllable resistive circuit independent of one anotherbased on the measurement voltage.

According to a first aspect of embodiments described below, an efficientand compact device is provided that is configured to apply a pluralityof voltage and resistance values to a control terminal of asemiconductor-based switch with a switch-on path or activation path forswitching on (activating) a semiconductor-based switch. The plurality ofvoltage and resistance values provides an advantageous concept forexactly controlling the semiconductor-based switch. Using a switch-onpath allows compact realization of the concept. It is an advantage ofthis concept that by using a switching path (and hence a reducednumber), a plurality of voltage values and/or resistance values can beset and the transmission parameter of the path can be changeddynamically.

A device for switching a semiconductor-based switch according to thefirst aspect includes a controllable activation voltage source that isconfigured to provide a time-varying voltage potential. The controllableactivation voltage source is connected to a controllable resistivecircuit in a series connection. The controllable resistive circuitcomprises at least two ohmic resistances connected in parallelcontrolled such that at least three resistance values of the parallelconnection result. A control device of the device is configured tocontrol the controllable activation voltage source and the controllableresistive circuit independent of one another, such that based ontime-varying resistance values and time-varying voltages, a time-varyingvoltage potential is obtained that is applied to a control terminal of apower switch via a circuit path.

According to a second aspect of embodiments described below, anefficiency of a gate driver is increased in that charge carriers removedfrom a control capacitance, such as an insulated gate of an IGBT, duringa switching process are used for operating the control device and arehence recycled. It is an advantage of the second aspect that by usingthe charge carriers for operating the control device, switching lossesare reduced and the operation can be made more energy-efficient.

A device according to the second aspect includes a terminal that isconfigured to be connected to a control terminal such as a gate terminalof a semiconductor-based switch. Further, the device comprises acontrollable deactivation voltage source that is configured to provide,at least temporarily a switching potential at a potential node connectedto the terminal and a control device that is configured to control thecontrollable deactivation voltage source accordingly. The switchingpotential is galvanically coupled to a supply node to which a supplypotential of the control device is coupled and has a lower potentialvalue than a threshold voltage of the semiconductor-based switch. Thecontrol device is configured to control the controllable deactivationvoltage source such that, based on the provided switching potential,charge carriers flow out of the control capacitance and contribute to anoperation of the control device based on the galvanic coupling.

According to a third aspect of embodiments described below, a conceptfor contactless detection of a switching state of thesemiconductor-based switch is provided, which provides precise detectionof a current change velocity of a power current flowing through asemiconductor-based switch.

An electric circuit according to third aspect includes asemiconductor-based switch with a first terminal and a second terminalfor conducting a power current and a sensor for detecting a currentchange velocity of the power current. The sensor includes an insulatingfoil that is configured to be connected to the first or second terminalof the semiconductor-based switch and an inductance arranged on theinsulating foil on a side of the same that is arranged opposite to aside facing the semiconductor-based switch during a measurementoperation of the sensor. The inductance is configured to detect themagnetic field generated by the power current and to provide ameasurement voltage based on the detected magnetic field. The inductanceis spaced apart from the semiconductor-based switch during themeasurement operation at least by the insulating foil, such thatcontactless measurement of the current change velocity by the sensor isenabled. The insulating foil comprises a mounting portion that isconnected to the first or second terminal of the semiconductor-basedswitch.

The insulating foil allows contactless detection of the power currentsuch that high voltage safety is obtained. Further, the detection isvery accurate since an inductance value of the inductance is exactlydetermined. Further, by means of an adjustable distance and/or anadjustable orientation between inductance and current-carrying elements,a measurement voltage of the sensor can be precisely adjusted.

Embodiments of the invention will be discussed in more detail below. Inthe figures, the same or equal elements are provided with the samereference numbers.

Embodiments of the three aspects can be combined with another and allowa mutual advantageous further development of the respective aspect.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will be detailed subsequentlyreferring to the appended drawings, in which:

FIG. 1 is a schematic block diagram of a device for switching asemiconductor-based switch according to the first aspect;

FIG. 2 is a schematic block diagram of a device for switching thesemiconductor-based switch according to the second aspect;

FIG. 3 is a schematic perspective view of an electric circuit with thesemiconductor-based switch and a sensor for detecting a current changevelocity of a power current flowing through the semiconductor-basedswitch according to the third aspect;

FIG. 4a is a schematic top view of the sensor, where an insulating foilincludes a mounting portion comprising an open recess;

FIG. 4b is a schematic top view of the sensor, where a mounting portionincludes a recess implemented in the form of a hole (closed recess);

FIG. 4c is a schematic top view of the sensor, where the insulating foilcomprises a mounting portion and where the current conductor is part ofintermediate circuit rails;

FIGS. 5a-5b are schematic block diagrams of a device comprising acombination of the first, second and third aspect and configured tocontrol the semiconductor-based switch;

FIG. 6 is a schematic illustration of the semiconductor-based switchwhere, a first sensor for detecting the current change velocity of thepower current is arranged at the first power terminal and wherein asecond sensor for detecting the current change velocity of the powercurrent is arranged at the second power terminal;

FIG. 7 is a schematic block diagram of an electric circuit comprising acommutation circuit with a first voltage potential and a second voltagepotential;

FIG. 8 is schematic diagrams of current or voltage curves at thesemiconductor-based switch during an activation process;

FIG. 9 is the voltage curves as well as the current curve of FIG. 8 fora deactivation process, for example a switch-off process;

FIG. 10 is a standard topology of a GDU with its functional blocksaccording to conventional technology;

FIG. 11 is a basic structure of a single-stage/multi-stageanalog-controlled GDU with an active change of the gate droppingresistor according to conventional technology;

FIG. 12 is a basic structure of a GDU having two voltage levelsaccording to conventional technology;

FIG. 13 is the basic structure of an single-stage/multi-stageanalog-controlled GDU according to conventional technology;

FIG. 14 is an embodiment of a single-stage/multi-stagedigital-controlled GDU according to conventional technology; and

FIG. 15 is a further embodiment of a digital control according toconventional technology.

DETAILED DESCRIPTION OF THE INVENTION

Before embodiments of the present invention will be discussed in moredetail below based on the drawings, it should be noted that identical,functionally equal or equal elements, objects and/or structures in thedifferent figures are provided with the same reference numbers, suchthat the description of these elements illustrated in the differentembodiments is inter-exchangeable or inter-applicable.

Embodiments describe below relate to switching a semiconductor-basedswitch, such as a metal oxide semiconductor field effect transistor(MOSFET) or an insulated gate bipolar transistor (IGBT).

The semiconductor-based switch can be of an n channel or p channel type.Further, the semiconductor-based switch can have a normally conductiveor normally non-conductive configuration. Based on a concentration ofcharge carriers stored on a control capacitance (gate capacitance)connected to the control terminal of the semiconductor-based switch(gate terminal), a switch state, for example conductive ornon-conductive, can be changed.

Thus, the control terminal of the semiconductor-based switch can be, forexample, a gate terminal of a MOSFET. Alternatively, the controlterminal can be the terminal of an insulated gate electrode of an IGBTor a different controllable semiconductor-based switch.

For improved clarity, the semiconductor-based switch will be describedbelow such that the semiconductor-based switch is in a non-conductivestate (i.e., non-conducting or only slightly conducting) when theconcentration of charge carriers in the control capacitance is so lowthat a control potential or a control voltage at the control terminal isbelow a threshold value of the switch (deactivated state). If theconcentration of charge carriers has a higher value, such that thecontrol voltage is above the threshold, the semiconductor-based switchis in a conductive state (activated state). It is obvious that thesestates can be interchanged based on the configuration of thesemiconductor-based switch, for example normally conductive/normallynon-conductive. When dynamically switching the switch, a Miller-inducedre-bias (reactivation) of the semiconductor-based switch can occur, suchthat in this case a change to the activated state is enabled when acontrol voltage is greater than a Miller voltage of thesemiconductor-based switch that is greater than the threshold voltage.Changing into the deactivated state is enabled when the control voltageis lower than the threshold voltage.

Supplying charge carries to the control capacitance for switching asemiconductor-based switch into an active state will be referred to asactivation below. Discharging charge carriers from the controlcapacitance for switching the semiconductor-based switch to thedeactivated state is referred to as deactivation. When thesemiconductor-based switch is in an inhibited, i.e., non-conductivestate, the same can be converted i.e. switched to the conductive ornon-conductive state by supplying or discharging charge carriers to orfrom the control terminal.

FIG. 1 shows a schematic block diagram of a device 10 for switching asemiconductor-based switch 12 according to the first aspect. The device10 includes a terminal 14 that is configured to be connected to acontrol terminal of the semiconductor-based switch 12 and a controllableactivation voltage source 16 that is configured to provide electricvoltage and charge carriers resulting, in a control capacitance of thesemiconductor-based switch 12, in a control voltage and an active stateof the semiconductor-based switch when the control voltage is greaterthan or equal to a threshold voltage of the same. The device 10 includesa controllable resistive circuit 18 including at least two ohmicresistances k1,1 and k1,2 connected in parallel. Based on the parallelconnection, the two ohmic resistances k1,1 and k1,2 can be switched suchthat four resistance values of the controllable resistive circuitresult. For this, the controllable resistive circuit 18 has acontrollable switch 22 a that is connected in series into a circuit pathwith the ohmic resistance k1,1. The series connection is connected tothe terminal 14 and forms an activation path. Further, the controllableresistive circuit 18 has a second controllable switch 22 b which isconnected in series into a circuit path with the ohmic resistance k1,2.The two circuit paths are connected into a parallel connection.

In a first state of the controllable resistive circuit 18, for example,the controllable switches 22 a and 22 b are each in a closed state, sothat both ohmic resistances k1,1 and k1,2 are effective in thecontrollable resistive circuit 18. In a second or third state, forexample, either the controllable switch 22 a or the controllable switch22 b is in the closed state and the respective other controllable switch22 b or 22 a in an open state, such that either the ohmic resistancek1,1 or k1,2 is effective in the controllable resistive circuit 18. In afourth state of the controllable resistive circuit 18, the controllableswitches 22 a and 22 b are, for example, in an open state, such thatboth circuit paths are interrupted and the controllable resistivecircuit 18 has a high, up to possible infinitely high, resistance value,i.e., the ohmic resistances k1,1 and k1, 2 are ineffective.

The device 10 includes a control device 24 that is configured to controlthe controllable activator voltage source 16 and the controllableresistive circuit 18 independent of one another in a time-varyingmanner. The control device 24 is, for example, configured to set statesof the controllable switches 22 a and/or 22 b in a time-varying manner.The controllable switches 22 a and/or 22 b can be implemented astransistors.

This allows that starting from the controllable activation voltagesource 16 up to the terminal 14 merely one path can be implemented(wired) on a circuit, for example a printed circuit board in order toobtain the functionality of variable voltages and variable resistancevalues. This means that implementation (wiring) of several paths betweenwhich switching is performed can be prevented and merely one path can bearranged which is used at all times.

The independent control of the controllable activation voltage source 16and the controllable resistive circuit 18 allows a high degree offreedom when configuring a switch on or switch-off path since, ascompared to hard-wired paths, both the voltage and the effective ohmicresistance can be varied independent of one another.

Further, the control device 24 is configured to set an output voltage ofthe controllable activation voltage source 16. The output voltage of thecontrollable activation voltage source 16 can, for example, include avoltage range of greater than or equal to −100 V and less than or equalto +100 V, greater than or equal to −10 V and less than or equal to +10V or a range of greater than or equal to 0 V and less than or equal to 5V.

The controllable activation voltage source is connected in series to thecontrollable resistive circuit 18 into a series connection. Based on thetime-varying control of the controllable activation voltage source 16and time-varying resistance values of the controllable resistive circuit18, a time-varying voltage potential can be applied to a controlterminal of the semiconductor-based switch 12. This means that thetime-varying voltage potential can be varied in that the control device24 changes an output voltage of the controllable activation voltagesource 16 in a time-varying manner and/or changes i.e., controls theresistance value of the controllable resistive circuit 18 in atime-varying manner.

An amount of charge carriers that are supplied per time unit to thecontrol terminal or away from the control terminal and hence a switchingvelocity of the semiconductor switch 12 can be set based on the voltageof the controllable activation voltage source 16 and based on theresistances k1,1 and k1,2.

Alternatively, the controllable resistive circuit 18 can also beconfigured such that one of the ohmic resistances k1,1 or k1,2 iseffective and thus a first or second state of the resistance values canbe set. Based on a control of the respective different switch 22 band/or 22 a, a third state of the controllable resistive circuit 18 canbe set, this means the two ohmic resistances k1,1 and k1,2 connected inparallel can be controlled such that three resistance values result.Alternatively, the third state can also be obtained when both ohmicresistances k1,1 and k1,2 are inactive. Alternatively, the controllableswitches 22 a and 22 b could also be controllable in the sense of logicOR-operation, such at least one of the ohmic resistances k1,1 or k1,2 orboth of them are effective (energized), such that the two ohmicresistances k1,1 and k1,2 can be controlled such that three resistancevalues result. One of the resistance values of the ohmic resistancesk1,1 or k1,2 can be configured, concerning its resistance value, withregard to a desired current change velocity (di/dt) during the switchingphase. The respective different ohmic resistance k1,2 or k1,1 can beconfigured, with respect to its resistance value with regard to adesired voltage change (du/dt) at a load path of the semiconductor-basedswitch 12. The voltage change relates, for example, to a changed voltagedrop between a collector and an emitter terminal (IGBT) or a changedvoltage drop between a source and a drain terminal (MOSFET). The tworesistances k1,1 and k1,2 can have the same or a differing value.

Alternatively, the controllable resistive circuit 18 can also comprisemore than two ohmic resistances that are controlled such that three,four or more resistance values result.

While the device 10 has been described such that a switch-on behavior ofthe semiconductor-based switch 12 is controllable by means of thecontrollable activation voltage source 16 and the controllable resistivecircuit 18, the device 10 can also be used for controlling a switch-offbehavior of the semiconductor-based switch 12. The controllableactivation voltage source 16 can, for example, be controlled as voltagesink, for example in that the provided voltage is lower than thethreshold voltage of the semiconductor-based switch 12.

This allows advantageous control of the semiconductor-based switchduring a switch-on and/or switch-off phase of the semiconductor-basedswitch 12.

A controllable voltage source in connection with a controllableresistive circuit can additionally also be arranged in a deactivationpath in order to allow exact and interference-free deactivation of thesemiconductor-based switch 12. In other words, the first aspect can alsobe used for deactivation.

FIG. 2 shows a schematic block diagram of a device 20 for switching thesemiconductor-based switch 12 according to the second aspect. The device20 includes the terminal 14 and a controllable deactivation voltagesource 26 connected to the terminal 14 and configured to provide atleast temporarily, i.e., in a time-varying manner, a switching potentialV_(DD) and a potential based on the switching potential V_(DD),respectively, at a potential node 27. The controllable deactivationvoltage source 26 forms a deactivation path.

The controllable deactivation voltage source 26 includes a controllableimpedance 27 and an output resistance Roff. The controllable impedance25 is configured to induce a variable control current I_(var).Alternatively, the controllable impedance 25 can also be an outputimpedance of the deactivation voltage source 26. The deactivationvoltage source 26 can be configured to apply a variable output currentI_(var) to the output resistance Roff of the controllable deactivationvoltage source 26. Alternatively or additionally, the provided voltagecan also be controllable and hence variable.

The controllable deactivation voltage source 26 includes a controllableswitch 33 connected between the controllable impedance 25 and the supplynode 29, such that, when the controllable switch 33 is in a closedstate, galvanic coupling exists between a supply node 29 and thecontrollable impedance 25.

The device 20 includes a control device 28 that is configured to controlthe controllable deactivation voltage source 26 in a time-varyingmanner. Based on the control of the control device 28, the controllabledeactivation voltage source 26 is configured to provide the switchingpotential V_(DD) and a potential based on the switching potentialV_(DD), respectively, during a switching interval during which thesemiconductor-based switch 12 is deactivated. The switching potentialallows a discharge of charge carriers from the control capacitance ofthe semiconductor-based switch 12, such that the same is converted inthe deactivated state. The switching potential V_(DD) has a potentialvalue that is lower than the threshold voltage of thesemiconductor-based switch 12.

The switching potential V_(DD) is galvanically coupled to the supplynode 29 to which a supply potential U_(S) of the control device 28 isapplied when the controllable switch 33 is closed. This enables that,when the semiconductor-based switch 12 is in the active state, i.e.charge carriers are stored on or in the control capacitance and theswitching potential V_(DD) is applied to the terminal 14, the chargecarriers can be discharged at least partly from the control capacitancein the direction of the switching potential V_(DD).

Based on the galvanic coupling of the supply potential U_(S) of thecontrol device 28 with the switching potential V_(DD), these chargecarriers contribute to the operation of the control device 28. Thecharge carriers being discharged from the control capacitance during aswitching process of the semiconductor-based switch 12 (deactivation)are used again or further which can also be referred to as recycling.Such a recovery of the control energy during switching-off allowsreduction of the energy consumption of the control device 28 and hencethe device 20. The reduced energy consumption results in an increasedefficiency of the device 20.

Alternatively or additionally, the device 20 can additionally comprise acontrollable resistive circuit, such as the controllable resistivecircuit 18. The controllable resistive circuit can be connected to thecontrollable activation voltage source 29, for example in a seriesconnection. The control device 28 can be configured to control thecontrollable resistive circuit. This allows exact control of theswitching behavior of the semiconductor-based switch 12 duringdeactivation and activation. Further, the supply potential U_(S) can beapplied to further active circuit elements or circuit groups of thedevice 20, such as further circuits or amplification circuits. The samecan be referred to as low voltage (LV) periphery, whereas current pathsthrough which the power current flows can be referred to as high voltage(HV) components. In this way, the recycled charge carriers can alsocontribute to an operation of the LV periphery.

A combination of the second aspect with the first aspect can beperformed, for example such that the second aspect, such as device 20,also comprises a controllable resistive circuit. The controllablevoltage source can be realized, for example, by means of a constantvoltage source or a potential that is connected to a controllableimpedance in order to obtain a variable voltage value at thecontrollable impedance, such that the controllable impedance and thepotential at least partly form the functionality of the controllablevoltage source, wherein the controllable voltage source can includefurther components.

This can be performed as an alternative to using the first aspect asdescribed with regard to FIG. 1.

FIG. 3 shows a schematic perspective view of an electric circuit 30 withthe semiconductor-based switch 12 and a sensor 31 for detecting acurrent change velocity of a power current I flowing through thesemiconductor-based switch 12 according to the third aspect. The sensor31 includes an insulating foil 32. An inductance 34 is arranged on theinsulating foil 32. The semiconductor-based switch 12 comprises a firstpower terminal 36 a and a second power terminal 36 b. The first powerterminal 36 a can, for example, be a drain terminal or a sourceterminal, when the semiconductor-based switch 12 is a MOSFET basedswitch. Alternatively, the first power terminal 36 a can be a collectorterminal or an emitter terminal when the semiconductor-based switch 12is an insulated gate bipolar transistor. The second power terminal 36can be the other terminal source and drain, respectively or emitter andcollector, respectively. Alternatively, the power terminal 36 a and/or36 b can also be arranged in the semiconductor-based switch and can beconnected to a current conductor allowing a connection to a powersupply.

If the power current I flows through the semiconductor-based switch 12,for example from the first power terminal 36 a to the second powerterminal 36 b, the power current I will generate a magnetic field 38.The inductance 34 is configured to detect the magnetic field 38 in orderto provide a measurement potential 42 based on a current change velocitydl/dt.

The inductance 34 is arranged on a side of the insulating foil 32arranged facing away from the semiconductor-based switch 12 and a powerterminal 36 a or 36 b of the semiconductor-based switch 12,respectively. This means that the inductance 34 is spaced apart from thesemiconductor-based switch 12 at least by the insulating foil 32.Further spacing sheets that can include insulating materials can bearranged between the insulating foil 32 and the semiconductor-basedswitch 12. The insulating foil 32 as well as the optional spacing sheetsresult in electrical insulation of the inductance 34 with respect to thesemiconductor-based switch 12. A distance between the inductance 34 andthe semiconductor-based switch 12 increased due to the spacing sheetscan hence result in an increased electrical insulation. This has theeffect that detecting the magnetic field 38 and hence detecting thecurrent change velocity dl/dt can be performed in a contactless manner,this means that a direct, for example galvanic coupling between thesemiconductor-based switch 12 and the sensor 31 can be prevented. Thisresults in an increased voltage security with respect to a measurementof a current change velocity based on leakage inductances arrangedwithin the semiconductor-based switch 12. Further, the inductance 34 canhave a high accuracy of the inductance value caused by a productionprocess. This enables the omission of using leakage inductances fordetecting the current change velocity in the semiconductor-based switch12. Alternatively or additionally, by means of the spacing adjustable bythe insulating foil 32 and possible spacing sheets, the amplitude of themeasurement voltage 42 can be adjusted. Increasing the distance can beused for reducing the measurement voltage. Alternatively, by means ofreducing the distance, for example by means of a lower foil thickness,the amplitude can be increased.

The mounting portion 45 can include, for example, a recess of theinsulating foil 32. The recess can be configured such that, for example,when the first and/or second power terminal 36 a and/or 36 b areimplemented as screw connection for connecting with a screw 46, theinsulating foil 32 can be arranged between the semiconductor-basedswitch 12 and the screw 46 without necessitating removal of the screw 46from the power terminal 36 a and 36 b, respectively. This can beperformed, for example, by inserting the insulating foil 32 between thescrew 46 and the semiconductor-based switch 12. A current terminal 47can also be arranged between the screw 46 and the semiconductor-basedswitch 12, for example, in the form of a cable lug or other suitablemeans, such that when mounting the screw 46 with respect to the powerterminal 36 a and 36 b, respectively, apart from an electrical, also amechanical fixing of the sensor 38 and the current conductor 47 isobtained. Alternatively, the current conductor 47 can also be guided ina direction to the power terminal 36 a having an angle to a surfacenormal of an area of the semiconductor-based switch 12 on which thepower terminal 36 a is arranged. The angle can be smaller than or equalto 90°. Simply put, the current conductor 47 can be led to thesemiconductor-based switch 12, for example in terms of a busbarterminal. The insulating foil 32 can be arranged parallel (for exampledirectly adjacent or arranged thereon) to the current conductor 47. Thisallows increased magnetomotive force of the inductance 34. Alternativelyor additionally, the power terminal 36 a and allocation, respectively,where the same is connected to the current conductor 47 can also have aspacing from the semiconductor-based switch, such that the magneticfield 38 has a homogeneous shape at the location of the inductance 34.

The mounting portion 45 is configured at least partly to surround thecurrent conductor, for example, in the form of a section of the powerterminal or a current conductor arranged thereon.

The inductance 34 has at least one preferential direction 44. Thepreferential direction 44 can, for example, be an axial direction alongwhich turns of a coil which at least partly form the inductance 34 arearranged. A sensitivity of the inductance 34 with respect to themagnetic field 38 ca be adjusted based on an orientation of thepreferential direction 44 with respect to direction of the current flowI. In this way, the sensitivity can be high or even at a maximum whenthe preferential direction 44 is arranged perpendicular to the currentflow direction 44. Alternatively, when the preferential direction 44 hasa different angle to the current flow direction, a sensitivity can belower than a maximum in order to prevent overdrive, i.e., a high or toohigh signal level of the measurement voltage 42.

The inductance 34 enables, for example by inductance, detection of acurrent change velocity of the current I. Such a detection can be highlyprecise compared to usage of parasitic leakage inductances in thesemiconductor-based switch 12. While conventionally used leakageinductances are a possibly non-specified characteristic of thesemiconductor-based switch 12, the inductance 34 can have low parametervariations, such that the measurement potential 42 enables preciseevaluation of the current change velocity. The precise evaluation allowsprecise monitoring and/or setting a state of the semiconductor-basedswitch 12, for example during a switching process.

The preferential direction 44 can have a tolerance range, for exampledue to production reasons, where the preferential direction 44 isperpendicular to the direction of the current flow I. The tolerancerange can have, for example, a value of less than or equal to ±15°, lessthan or equal to ±10° or less than or equal to ±5°. Alternatively, theinductance 34 can also have a second preferential direction, for examplealong a direction that is opposite to the preferential direction 44, forexample when the inductance 34 is a symmetric coil.

The inductance 34 can have a flow concentrator that is configured toconcentrate a magnetic flow of the magnetic field. The flow concentratorcan, for example, be a ferrite core that is arranged as coil core withregard to the inductance 34.

The measurement voltage 42 can be transmitted, for example, via anelectrically shielded transmission line, for example to a controldevice, for example the control device 24 or 28.

Alternatively, the inductance can also be a series connection ofinductive elements, such as a series connection of several coils. Thecoils can have, when the same comprise a flow concentrator, a common orseparate flow concentrators. The above-described aspects can be combinedwith one another. For example, the first aspect can be combined with thesecond aspect and/or the third aspect. Alternatively or additionally,the second aspect can be combined with the third aspect and/or the firstaspect.

Alternatively, the first and/or second power terminal 36 a and/or 36 bconfigured to be connected to the current conductor 47 by means of aplug connection, a clamp connection, a solder connection or othermechanical means. Accordingly, the current conductor 47 can also beconfigured, as an alternative to the implementation with a cable lug,according to the respective mechanical connection.

The measurement voltage can be transmitted via a measurement line 49,for example a single-wire or two-wire line. The measurement line 49 cancomprise a shielding that is configured to reduce electromagneticspurious influences of the measurement voltage 42 and/or with respect tothe measurement voltage 42. Alternatively or additionally, themeasurement voltage can be transmitted via a twisted two-wire line.

FIG. 4a shows a schematic top view of the sensor 31, wherein theinsulating foil 32 includes a mounting portion 45 a comprising an openrecess (open mounting portion). The recess enables that the insulatingfoil 32 can be arranged on the semiconductor-based switch withoutremoving possible mounting screws. The inductance 34 has thepreferential direction 44 that is arranged in an orientation parallel tothe mounting portion 45 a. The mounting portion 45 a partly surroundsthe current conductor 47.

FIG. 4b shows a schematic top view of the sensor 31, wherein a mountingportion 45 b includes a recess implemented in the form of a hole (closedrecess). The closed recess and the hole, respectively, can have a round,rectangular, elliptic or free-form peripheral geometry. The closedrecess allows guiding of a current conductor through the mountingportion 45 b, for example when the current conductor 47 is guided to thesemiconductor-based switch 12 via a plug connection. If the currentconductor 47 is guided through the mounting portion 45 b, the mountingportion 45 b surrounds the current conductor 47 completely. Theinsulating foil 32 comprises connecting terminals 166 a and 166 b thatare connected to the inductance 34 and configured to provide, duringoperation of the sensor 31, a measurement voltage and a measurementpotential to the inductance 34. The measurement line 49 can be connectedto the connecting terminals 166 and 166 b, for example by a mechanicallyfixed connection as it can be obtained by soldering or screwing. Theinductance 34 has the preferential direction 44 which is rotated withrespect to the illustration in FIG. 4a , such that the preferentialdirection 44 is arranged in a direction (or 180° offset thereto) towardsthe mounting portion 45 b.

FIG. 4c shows a schematic top view of the sensor 31 comprising amounting portion 45 c which is implemented as closed recess. The currentconductor 47 is part of intermediate circuit rails 51 that areconfigured to supply the semiconductor-based switch 12 with electricenergy or receive it from the same. The sensor 31 comprises the twoinductances 34 a and 34 b that are arranged perpendicular to a directionof the current flow I with the preferential direction 44.

In the following a combination of the first aspect, the second aspectand the third aspect will be described.

FIGS. 5a and 5b show schematic block diagrams of a device 50 that isconfigured to control the semiconductor-based switch 12.

The device 50 comprises a controllable switch-on path (activation path)48, a controllable switch-off path (deactivation path) 52, the sensor 31for detecting the current change velocity of the power current flowingthrough the semiconductor-based switch 12 and a control device 56. Thecontrol device 56 is configured to control the activation path 48 duringan activation phase of the semiconductor-based switch 12. During adeactivation phase, the control device 56 is configured to control thedeactivation path 52. As discussed in detail below, the activation path48 includes features of the first aspect. The deactivation path 52comprises features of the first and second aspects. The sensor 31comprises features of the third aspect.

The activation path 48 includes a controllable activation voltage source58 referred to as functional unit “VonMid-Control”. The controllableactivation voltage source 58 comprises a digital to analog converter(DAC) 62 that is configured to receive a time-varying digital signalVonMidRef from the control device 56, and to convert the same into atime-varying analog signal. The controllable activation voltage source58 further comprises a differential amplifier 64 that is configured tocompare the time-varying analog signal provided by the DAC 62 withrespect to a source voltage V_(CC) and to amplify a difference of thetwo signals VonMidRef (and its analog representation, respectively) andV_(CC) and in this way to provide a time-varying voltage VonMid. Thecontrollable activation voltage source 58 is configured to provide atime-varying output voltage based on the time-varying difference.Alternatively, for example, the DAC 62 can be configured to detune avoltage regulator. The same is configured to provide the desired voltageVonMid fed from the source voltage V_(CC) (and hence the maximumpossible voltage).

The activation path 48 includes a controllable switch device“Von-control” 66 comprising two controllable switches 68 a and 68 b aswell as a multiplexer “VonMux” 72. The multiplexer 72 is configured, forexample, to receive control instructions from the control device 56, tomultiplex the same and to control the controllable switches 68 a and 68b based on the multiplexed control signals. Alternatively, thecontrollable switch device 66 can also be configured to obtain a controlsignal with respect to one or all controllable switches 68 a/68 b. Onthe input side, the controllable switch 68 a is connected to thecontrollable activation voltage source 58. On the input side, thecontrollable switch 68 b is connected to the source potential V_(CC). Onthe output side the two controllable switches 68 a and 68 b areconnected to one another and are connected to a controllable resistivecircuit 74 via a common potential connection. When the controllableswitch 68 a is in a closed state and the controllable switch 68 b is inan open state, the variable voltage VonMid, and when the controllableswitch 68 a is in the open state and the controllable switch 68 b is inthe closed state, the source voltage V_(CC) can be applied to thecontrollable resistive circuit 74. The source voltage can, for example,have an almost constant value of, for example, approximately 5 V or 10 V(for example, for MOSFET-based switches) or 15 V (for example, forIGBT-based switches). Basically, the source voltage V_(CC) cancorrespond to the voltage by which the respective semiconductor-basedswitch is fed statically in the on-state and, hence, can also have adifferent value.

The controllable resistive circuit 74 includes the ohmic resistancesk1,1 and k1,2 that are connected in parallel to one another as describedfor the controllable resistive circuit 18. Further, the controllableresistive circuit 74 comprises a multiplexer 76 that is configured toreceive control signals from the control device 56 to control, based onthe received control signals, controllable switches 57 a and 57 b of thecontrollable resistive circuit 74 as described for the controllableresistive circuit 18. Alternatively, the controllable resistive circuit74 can also be configured to obtain, with respect to one or allcontrollable switches of the same, a control signal, i.e., themultiplexer is an optional element.

On the output side, the activation path 48 is connected to the terminal14, wherein an ohmic resistance R_(S) is connected between a node point78 and the terminal 14. The activation path 48 is configured to converti.e., to switch the semiconductor-based switch 12 from a deactivatedstate to an activated state (activation), based on charge carriersprovided by the controllable activation voltage source 58 or the sourcevoltage V_(CC) and entering the control capacitance.

The deactivation path 52 is configured to receive, in the function of avoltage sink, charge carriers from the control capacitance of thesemiconductor-based switch 12 (discharge the same) and in this wayconvert the semiconductor-based switch 12 in a deactivated state. Thiscan be performed such that during the deactivated state, i.e., at an endof the switching process, a holding voltage, i.e., a static OFF voltageV_(EE) is applied to the control terminal. The voltage V_(EE) can have anegative potential with respect to a reference voltage, such as theemitter potential of the semiconductor-based switch 12 and/or a groundvoltage (GND), such that, based on the negative potential, the controlterminal can be discharged and/or biased with a negative potential.

The deactivation path 52 comprises a series connection of a controllableresistive circuit 82 and a controllable (deactivation) voltage source(Voff-Control) 84 that is configured to provide the switching potentialV_(DD) at the potential node 27. The deactivation path 52 is connectedto the node 78 and hence to the terminal 14. The controllable voltagesource 84 comprises two controllable switches 86 a and 86 b. At oneterminal, the controllable switch 86 b is connected to the static OFFvoltage V_(EE). The controllable switch 86 a is connected to theswitching potential V_(DD), wherein a controllable impedance 88 isconnected between the controllable switch 86 a and the switchingpotential V_(DD), which is configured to set, based on controlinstructions of the control device 56, a time-varying impedance betweenthe controllable switch 86 a and the switching potential V_(DD).

The switching potential V_(DD) is galvanically connected to a supplypotential of the control device 56, such that charge carriers flowingfrom the control capacitance of the semiconductor-based switch 12 viathe controllable resistive circuit 82 and via the controllable switch 86a to the switching potential V_(DD) contribute to the operation of thecontrol device 56. The static OFF voltage V_(EE) can have a voltagevalue that is lower than the threshold voltage of thesemiconductor-based switch 12 and/or can have a voltage value that islower than a reference voltage, for example ground (GND), such thatactivation of the semiconductor-based switch 12 is prevented when thestatic OFF voltage V_(EE) is applied to the terminal 14 via thecontrollable resistive circuit 82 and the controllable switch 86 b. Inother words, with respect to GND, the static OFF voltage V_(EE) can benegative or equal to GND and can have any value between a maximumemitter voltage and 0 V.

The controllable voltage source 84 is configured to receive controlinstructions from a sequence control of the control device 56. Based ona control of the controllable switches 86 a and 86 b, the switchingpotential V_(DD) can be applied to the potential node 27 that isconnected to an input side of the controllable resistive circuit 82 whenthe controllable switch 86 a is closed. Alternatively, when thecontrollable switch 86 b is closed, the static OFF voltage V_(EE) can beapplied to the potential node 27 and to the controllable resistivecircuit 82, respectively. The two controllable switches 86 a and 86 bare connected to one another at the terminal facing away respectivelyfrom the voltage potential V_(DD) and V_(EE), respectively. Thepotential node 27 can be any point or portion along an electricconnection between the controllable deactivation voltage source 84 andthe controllable resistive circuit 82. Thus, a potential applied to thepotential node 27 can be switched in a time-varying manner duringdeactivation between the switching potential V_(DD) guided via acontrollable impedance (Ioff-Control) 88 and the static OFF-voltageV_(EE).

The controllable resistive circuit 82 comprises two ohmic resistancesk2.1 and k2.2 that are connected parallel to one another. A controllableswitch 22 c is connected between the ohmic resistance k2.1 and thecontrollable voltage source 84. A controllable switch 22 d is connectedbetween the ohmic resistance k2.2 and the variable voltage source 84,such that, based on a control of the controllable switches 22 c and 22d, at least three resistance values can be set, as it is described forthe controllable resistive circuit 18.

The controllable resistive circuit 82 comprises a multiplexer 92 that isconfigured to receive control instructions for controlling thecontrollable switches 22 c and 22 d from the control device 56.Alternatively, the controllable resistive circuit 82 can also beconfigured to obtain a control signal with respect to one or allcontrollable switches 68 c/68 d.

The sensor 31 is configured to provide the measurement voltage 42. Acomparator circuit 94 connected between the sensor 31 and the controldevice 56 is configured to receive the measurement voltage 42 and toprovide data signals to the control device 56 which comprisinginformation with regard to the current change velocity, for example whenswitching on (activating) the semiconductor-based switch 12, the currentchange in the active state as well as the current drop velocity whendeactivating the semiconductor-based switch. The control device 56 isconfigured to control the semiconductor-based switch 12 at least partlybased on the measurement voltage 42. For this, the comparator circuit 94can comprise two comparators 96 a and 96 b to which the measurementvoltage 42 can be applied, each to a first comparative terminal. Arespective second comparative terminal (input) can be connected to avariable reference potential. The variable reference potential can beprovided by multiplexers 98 a and 98 b, respectively, that can becontrolled by the control device 56. The respective data signals thatare based on the comparative results are provided to the control device56. Thus, the comparator circuit 94 is configured to compare anamplitude of the measurement voltage 42 with comparative values providedby the control device 56. The sensor 31 and/or the device 50 can furthercomprise a matching circuit for matching or adapting, i.e. scaling theamplitude of the measurement voltage 42, for example in the form of anattenuator.

The device 50 comprises an Active GateClamp 102 (active circuit at thecontrol terminal for preventing Miller-induced (re) switching-on of thesemiconductor-based switch 12) including a controllable switch, forexample a transistor 104 connected between the terminal 14 and thestatic off-voltage V_(EE), such when the controllable switch 104 isclosed, the terminal 14 and hence the control terminal of thesemiconductor-based switch 12 is connected to the static off-voltageV_(EE). This enables a discharge of charge carriers from the controlcapacitance and prevention of exceeding the threshold voltage.Alternatively, the transistor can also be connected to a referencepotential, such as OV or ground (GND).

The controllable switch 104 is configured to be controlled based on avoltage drop of a resistor Rs connected in parallel to the controllableswitch 104, wherein a lowpass filter (LPF) 106, which is configured tofilter short-term voltage drops, is arranged between the (sensor)resistor Rs and a control input of the controllable switch 104. Avoltage drop across the sample resistor Rs can result in closing of thecontrollable switch 104. In dependence on the dynamic voltage differencebetween the terminal 14 and the node 78, the path via the ActiveGateClamp 102 can have a high impedance or accordingly a low impedance.The stronger the deviation between a voltage at the terminal 14 and thenode 78, the lower the impedance of the Active GateClamp, i.e. the shuntpath 102, and hence counteracts a Miller-induced voltage rise. TheActive GateClamp is configured to be possibly effective only during thedeactivation process, such that switching on (activation) of thesemiconductor-based switch 12 is prevented at these times. This enablesthe avoidance of Miller-induced switching-on of the semiconductor-basedswitch 12. The lowpass filter 108 enables that merely shortdisturbances, such as by switching processes of the semiconductor-basedswitch 12 and the like are filtered out and in that way undesiredshort-circuiting of the control terminal is prevented. Further, thelowpass filter 106 and the sensor resistor 104, respectively, areconnected to a monitoring means (Vge/Vgs-Monitor) 122 that allowsdetection of the potential at terminal 14 by means of the control device56.

A first terminal of a controllable switch (ActiveOff) 108 is connectedto the terminal 14. A second terminal of the controllable switch 108 isconnected to the reference potential. The control device 56 isconfigured to control the controllable switch 108 such that based on thecontrol, the terminal 14 can be connected to the reference potential andthe static OFF-voltage V_(EE), respectively. For example, when thecontrol voltage of the semiconductor-based switch 12 falls below aconfigured threshold during deactivation, the control device 56 can beconfigured to control the controllable switch 108 to short-circuit thecontrol terminal. For example, a voltage level of the control voltagecan be defined as such a threshold, where the switch-off process(deactivation) is terminated. This allows reducing the control voltageby further discharging the control capacitance, for example, forincreasing the interference reserve of the semiconductor-based switch12. An interference can take place, for example, by a varying voltage atthe terminal 14 or the node 78, for example by electromagneticcoupling-in. At the same time, such a structure allows switching-off thesemiconductor-based switch to the static OFF-voltage V_(EE), such thatin this case switching off is enabled by bypassing the deactivation path52. In this case, the source voltage V_(CC) can be used as inversefeedback potential. Alternatively, the controllable switch 108 can alsobe connected to the reference potential GND.

Further, the device 50 includes a first inverse feedback branch 112(active di/dtoff-clamp) that is configured to limit the currentsteepness of the power current flowing through the switch 12 during thedeactivation phase. The inverse feedback branch is connected to themeasurement voltage 42 of the sensor 31 at a control input. An increasedcurrent drop velocity (current change velocity outside the controlrange) during a deactivation phase can result in a reduced potentialvalue of the measurement voltage 42. For a maximum allowable and/or adesired current drop velocity lying above the set amount of the controlrange in normal operation, a respective threshold of the measurementvoltage 42 can be determined. Based on the respective threshold, a Zenerdiode 114 can be designed, which is connected in the first inversefeedback path 112 between the measurement voltage 42 and a controlterminal of a transistor 115.

If the current drop velocity exceeds the threshold, the output voltageof di/dt sensor 31 falls below the limit. Thereby, the Zener diode 114becomes conductive and the inverse feedback becomes active by means of aconductive state of the transistor 115. This means that the switchingpotential V_(CC) is applied to the terminal 14 via an ohmic resistanceR. Thus, the current drop at the control terminal of thesemiconductor-based switch 12 can be reduced by feeding in a controlcurrent (gate current) of the active di/dtoff-clamp, such that thecurrent drop velocity falls. In other words, the di/dtoff-clamp consistsof a transistor with a resistor R connected with respect to the voltagesource V_(CC) of the driver output stage. The di/dt sensor 31 serves ascontrol source. Hence, the transistor 115 can be referred to as limitswitch for limiting the current change velocity during the deactivationprocess.

Further, the device 50 comprises a second inverse feedback branch(active di/dton-clamp) 116 that is configured to limit a current changevelocity, for example during the activation phase. During an error case,for example in the case of an error in the sequence control of thecontrol device 56 or error cases in the load circuit of the powerswitch, i.e. the semiconductor-based switch 12, resulting in currentchange velocities significantly above the defined set value, the inversefeedback branch 116 is configured to limit the current steepness asinverse feedback branch. Contrary to the normal control (control innormal case) this takes place in a lossy manner. The inverse feedbackbranch 116 includes a transistor 119 with ohmic resistance R connectedwith respect to the static OFF-voltage V_(EE) of the driver outputstage. The di/dt sensor 31 can be used as control source. If the currentchange velocity exceeds a threshold, the output voltage of the sensor 31rises above a limit based on which a Zener diode 118 is designed.Thereby, the Zener diode 118 and hence the transistor 119 becomesconductive and inverse feedback becomes active. In that case, the staticOFF-voltage can be used as inverse feedback potential. By dischargingpart of the gate current into the di/dton-clamp (inverse feedback branch116), the gate current change is reduced and the current change velocitydrops. Thus, the active di/dton-clamp can be referred to assuperordinate protective instance which is part of the security concept.

Further, the device 50 includes the voltage monitor (Vge/Vgs-Monitor)122 that is configured to detect the control voltage (gate voltage) andthe voltage at a terminal of the sensor resistor Rs facing away from thecontrol terminal, respectively and to compare the same with a thresholdprovided by the control device 56. The voltage monitor 122 is configuredto provide a differential signal based on the comparison and to providethe same to the control device 56.

For this, the voltage monitor 122 can comprise, for example a comparatorcircuit that is configured to compare the control voltage with a staticthreshold. The static threshold can, for example, be the thresholdvoltage of the semiconductor-based switch 12. The threshold voltage canbe variable during operation, i.e. at runtime. Thus, the voltage monitorcan be configured to check whether the gate threshold voltage isundershot. The control device 56 can be configured to control thecontrollable switch 108 based on the signal provided by the voltagemonitor 122. This enables prevention of undesirable (re-) bias, i.e.activation of the semiconductor-based switch 12 by a voltage rise viathe semiconductor-based switch 12. The voltage rise can increase thegate potential of the semiconductor-based switch 12 via the Millercapacitance.

In the following, reference is made to functions of the control device56 that are illustrated as functional blocks in a schematic blockdiagram. The control device 56 can, for example, be a digitallyprogrammed circuit, for example in the form of a digital signalprocessor (DSP), a programmable gate array (PGA) or in the form of amicroprocessor.

The control device 56 includes an input interface 124 referred to asRX-I F. The input interface 124 is configured to receive controlinstructions, for example from a superordinate control unit 126. Thesuperordinate control unit 126 can be galvanically separated from thedevice 50 and/or the control device 56. The control instructions can bereceived, for example as a pulse-width modulated (PWM) signal. Thecontrol instructions can be received via a receive channel, for exampleas a 1-bit data stream in order, for example, to receive ON/OFFinformation. Alternatively, the receive channel can also include symbolswith more than one bit.

The transmission and hence the reception of the control instructions atthe input interface 124 can be performed optically by means of anoptical interface. For that, the input interface can have the opticalinterface. This allows galvanic separation to the superordinateinterface 126. Further, optical data transmission allows superpositionof the data stream with additional information from the superordinatecontrol unit 126, for example, a main control of the digital-controlledactive gate drive unit (DCAGDU). The superimposed additional informationcan be modulated at runtime within the PWM-ON message as n-bit datastream. In that way, parameter adaptations, such as the currentintermediate circuit voltage to which the device 50 is connected canalso be transmitted. Alternatively, transmission can also be performedby means of electric signals.

Such information can be used by the control device 56 to adapt theswitch-on and switch-off sequence, respectively, in the range of thedu/dt phases. Other information for adaptation adapting the switchingsequences at runtime can also be transmitted. Further, the inputinterface 124 can be configured to implement a so-called watchdogfunction, i.e. a functionality check by the primary side (superordinatecontrol 126). This means that the superordinate control 126 can beconfigured to determine erroneous function of the control device 56 andthe transmission path, respectively, when the PWM signal is not or onlypartly received. The superimposed information can be implemented, forexample by an 8-bit value within the PWM signals. A realized protocolcan have, for example, two functions.

The first function can include control information. For this, thesuperordinate control 126 can transmit a primary-side switching commandON or OFF to the GCU. A second function can include a transmission of acurrent value of the intermediate circuit voltage (UZK) which can beapplied to the semiconductor-based switch during the on-command inreal-time. The same can be used by the GCU in order to determine thevoltage change velocity and adapt the same accordingly. Apart from that,implicit watchdog function of the primary side is implemented in that itis monitored whether the primary-side PWM pattern is permanentlygenerated.

The sequence is, for example, as follows:

The primary-side modem controller receives the switch-on signal of thecontrol as well as a digitized value for the height of the intermediatecircuit voltage. The respective PWM pattern is generated from thesevalues. The period duration of the PWM signal is identical and has to beshorter than the secondary-side short impulse suppression time constant.First, a static ON-signal is transmitted until the secondary-sidefeedback channel returns an ON-signal after passing through the shortimpulse suppression. From now on, a LOW-HIGH pattern with fixed periodduration is transmitted by the modem transmitter. The ratio of HIGH toLOW is proportional to the height of the intermediate circuit voltage.Accordingly, in an 8-bit digital value, the following limits result:

UZK=0->tlow/thigh=1/255; UZK=0.5*UZKmax->tlow/thigh=127/128;UZK=UZKmax->tlow/thigh=255/1.

Switch-off signaling is performed in dependence on the current PWMportion. If a HIGH portion is currently active, the modem willstatically change to LOW. Wth the confirmation of the feedback signalafter termination of the short impulse suppression, the modem sequenceis successfully terminated and allows new ON-commands.

If a LOW portion is currently active, the modem changes from LOW tostatic HIGH. The secondary-side feedback signal detects the static HIGHstate after termination of the short impulse suppression and changes toLOW. Thus, also the modem output statically returns to LOW and allowsnew ON-commands.

Further, the control device 56 comprises an output signal interface 128referred to as TX-IF. The output signal interface can be configured totransmit an active signal to the primary-side control instance, i.e. thesuperordinate control unit 126, for example when an error state, such asundervoltage, timeout, short-circuit or excessive temperature has beendetected. The control device 56 is configured to determine such an errorstate by means of internal monitoring.

Alternatively, the output signal interface 128 can be configured toimplement the respective signal transmission path towards thesuperordinate control unit 126 as watchdog of the secondary side (i.e.the device 50). For example, an alternating ON/OFF signal pattern can bepermanently transmitted from the output interface 128 in order tosignalize that the DCAGDU is active, i.e. the device 50 is operable. Inthe error case, a permanent error signal can be transmitted.Alternatively, the error signal can also be transmitted in intervals oronce. Alternatively, the output signal interface 128 can be configuredto transmit, in a deactivated state of the device 50, the signal with aconstant polarity and/or a constant intensity level, such that thesignal remains also static and constant, respectively.

Alternatively, the output signal interface 128 can be configured totransmit a feedback signal to the input signal interface 124. The outputsignal interface can be configured, for example to transmit a static“GOOD” signal when no PWM signal is applied to the input signalinterface 128, in order to signalize that no error has been detected. Inthe error case, the output signal interface 128 can be configured totransmit an error signal (or BAD signal). If a PWM ON-signal is receivedat the input signal interface 124, the feedback signal can betransmitted after the signal passage chain (signal processing) by theoutput signal interface 128 such that the respective signal comprises alevel change or polarity change in order to signalize that the PWMsignal has arrived and is being processed.

It is an advantage that the superordinate control unit 126 can beconfigured to determine, based on a time difference between a signaltransmitted to the device 50 and a signal received by the device 50based thereon, a signal runtime in real-time to the runtime and forexample in parallel connections of power switches, i.e. several devices10, 20 and/or 50 whether adaptations are to be performed forsynchronizing the PWM signals. Alternatively or additionally, during aPWM OFF-phase, i.e. while a command for deactivating thesemiconductor-based switch 12 is received, state information of thedriver side can be transmitted as serial data stream.

Further, the control device 56 includes a register database(TimingConfig Register) 132, for example in the form of a memory partlyor completely comprising control times for the dynamic switching phases.The control times can, for example, be used as basic configuration andas criterion for timeout, respectively, by the control device 56 forsequence control of switching processes. The control times can beprogrammed, i.e. stored as configuration data during programming thecontrol device 56. Alternatively or additionally, the control times canbe changed or replaced during operation, i.e. at runtime, for examplethe PWM protocol by the superordinate control unit 26 (primary-sidecontrol). The control device 56 is configured to readjust the controltimes of individual switching portions during switching processes,depending on the operating point, i.e. to perform independent adaptationof the control times.

The register database 132 is logically connected to the input signalinterface 124 and configured to receive information from the same.

The control device 56 includes a status register in the form of aregister databank 134. The status register 134 is coupled to the outputsignal interface 128 and configured to provide information to the same.

Status information of the device 50, i.e. the DCAGDU can be stored inthe status register 134. The same can, for example, be parameters fordetecting under-voltage errors, timeouts of the sequence control(protocol of the control device 56), overcurrent and/or short-circuiterrors. The control device 56 can be configured to return, via thefeedback path, i.e. the output signal interface 128, a respective statussignal to the primary state side control 126. Alternatively, the controldevice 56 can be configured to transmit detailed information to thesuperordinate control unit 126, for example in the form of a serialstatus data stream.

The control device 56 includes a functional unit for error handling(error handler) 136. The same is configured to set a secure state of thedevice 50 in the case of an overcurrent and/or a short-circuit and anunder-voltage, respectively. Acknowledging an error can, for example, beperformed by an “OFF”-signal that is transmitted via the PWM inputsignal and received at the input signal interface 124. Alternatively oradditionally, acknowledging can be performed such that, for example, theOFF-signal has to be applied for a defined time period. Such anacknowledging timeout (time duration of a signal level) can be freelydefined, for example more than 100 ns, more than 1 ms or more than 10 msor even longer.

The control device 56 includes a functional block (ConfigVge/Vgs-Monitor) 138 for configuring the Vge/Vgs-Monitor 122. The ConfigVge/Vgs-Monitor 138 is configured to provide the defined threshold tothe Vge/Vgs-Monitor 122, this means to transmit respective informationto the same. This can be in the form of a digital signal that can beconverted to an analog reference voltage Ref Vge,th by a DAC of theVge/Vgs-Monitor 122. The Vge/Vgs-Monitor 122 is configured to set, basedon the obtained signal, a specific threshold at a differential amplifieror a comparable circuit and to compare the gate voltage with regard toexceeding or falling below the threshold and to provide the exceeding orfalling below to a functional block for digital monitoring of the gatecontrol voltage (Sens Vge/Vgs-Monitor) 142.

The threshold can be generated, for example by switching-in a fixed orvariable reference voltage at the Vge/Vgs-Monitor 122. Alternatively,the threshold generation can be performed in a variable manner bygenerating a PWM duty cycle in combination with a connected lowpass andfunction of a DAC. Thereby, the analog value of the detected gatevoltage can be detected in discrete stages.

The control device 56 includes the functional block Sens Vge/Vgs-Monitor142 for digitally monitoring the gate control voltage. The same isconfigured to receive a signal transmitted by the Vge/Vgs-Monitor 122and to monitor the digital threshold. An information, for example basedon a comparison “Vgs/Vge<Ref Vge,th” can be provided to a functionalblock “ActiveOff Controller” 144. The ActiveOff Controller 144 isconfigured to control the controllable switch 108. Alternatively oradditionally, the Sens Vge/Vgs-Monitor 142 can be configured todetermine the threshold voltage from which the semiconductor-basedswitch 12 starts to carry current. This can be performed by means ofevaluating the output signals provided by the Vge/Vgs-Monitor 122 basedon the variable thresholds. Thus, the Sens Vge/Vgs-Monitor 142 can form,together with the Config Vge/Vgs-Monitor 122, an analog-to-digitalconverter (ADC).

Further, the control device 56 includes the ActiveOff Controller 144coupled to the functional module 142. The ActiveOff Controller 144 is anactuator module for controlling the controllable switch 108. If thecontrol device 56 detects that the gate voltage falls below the definedthreshold, the control device 56 can trigger activation of the ActiveOff108 (short circuit) which is directly attached, i.e. connected to thegate signal, i.e., the control terminal. Thereby, undesired (re-) biasof the semiconductor-based switch 12 can be prevented, for example bycoupling-in du/dt.

The control device 56 includes a configuration module Configdi/dt-Monitor 146, that is configured to configure the di/dt-Monitor 94,i.e., to provide comparative values to the same. The Configdi/dt-Monitor 146 is configured to set the multiplexers 98 a and 98 band to change the comparison thresholds of the comparators 96 a and 96b.

The di/dt-Monitor 94 is configured to detect or to monitor, in abidirectional manner, both the current change velocity duringswitching-on (activating) the semiconductor-based switch 12, the currentchange in the switched on state as well as the current drop velocityduring switching-off (deactivating) the semiconductor-based switch 12.The Config di/dt-Monitor 146 is configured to dynamically change thereference thresholds of the di/dt-Monitor 94.

The control device 56 further comprises a monitoring functional blockSens di/dt-Monitor 148 that is coupled to the Config di/dt-Monitor 146.The Sens di/dt-Monitor is configured to receive the comparative resultsof the di/dt-Monitor 94 and hence to monitor the current change velocity(di/dt) in two stages. The control device 56 is configured to detect, bymeans of the Sens di/dt-Monitor 148, a time instant at which the currentflow in the semiconductor-based switch 12 changes (beginning of thecurrent change), a short-circuit in the active state of thesemiconductor-based switch 12 as well as the set value of the currentchange velocity during activation (switching-on) and deactivation(switching off) of the semiconductor-based switch 12.

The control device 56 includes a functional block Config VonMid 152 thatis configured to control the DAC 62 of the controllable activationvoltage source 58 and to provide a digital signal to be converted to thesame, respectively. The functional block Config VonMid 152 can hence bedescribed as digital configuration of a control voltage for gradualadaptation of the current change velocity during the activation process.Based on gradual control of the DAC 62 for gradual adjustment of thecomparative voltage of the differential amplifier 64, thus, the voltageof the controllable activation voltage source 58 can also be adjusted. Aset value predetermined by the controllable activation voltage source 58can be determined by a threshold of the di/dt-Monitor 94 comparing a setstate with the current state. The functional block Config VonMid can beconfigured, for example, to change such a control voltage in 256 stages(8 Bit) in the range above the gate threshold voltage ad the staticgate-on-voltage and the source voltage V_(CC), respectively. This allowscontrol of the gate current and hence the current change velocity of thesemiconductor-based switch 12 with respect to the predetermined setvalue depending on the operating point. Alternatively, a variable outputvoltage of the activation voltage source 58 can also be obtained in thata constant voltage potential is connected to a variable outputresistance.

If the controllable activation voltage source 58 is, for example,configured to provide a constant voltage potential with variable outputresistors, the functional block Config VonMid 152 can be configured togradually adjust the output resistance, for example in 128, 256 or 512or more stages. Alternatively or additionally, when the controllableactivation voltage source 58 comprises both a gradually adjustableoutput voltage as well as a gradually adjustable output resistance, thefunctional block Config VonMid 152 can be configured to gradually changeboth the voltage potential and the output resistance.

Further, the control device 56 comprises a configuration block ConfigIoff 154 that is configured to adjust the controllable impedance 88.This can be performed, for example, by a digital configuration of thecontrollable impedance 88 (current sink) for gradual adaptation of thecurrent drop velocity to a definable set value during a deactivationprocess. The set value can be defined by a threshold of thedi/dt-Monitor 94. The functional block Config Ioff 154 is configured toset the controllable impedance 88 by means of a control voltage, forexample in 256 stages in the range below the static gate-on-voltage anda voltage above the gate threshold voltage. Alternatively, thefunctional block Config Ioff can be configured to vary the controlvoltage in one of 256 different numbers of stages, for example 128, 512,1024 or more. Thereby, a gate resistance (Rgoff) and hence the currentdrop velocity can be controlled to the set value depending on theoperating point.

Further, the control device 56 comprises a functional block for centralsequence control (FSM Closed Loop ON-Control/OFF-Control) 156. Thecentral sequence control 156 is configured to obtain information fromthe TimingConfig Register 132 and to update its information forobtaining information from the error handler 136 and to provideinformation to the status register 134 in order to control thefunctional block Config di/dt-Monitor 146 for obtaining information fromthe module Sens di/dt-Monitor 148 and to control functional blocksConfig VonMid 152 and Config Ioff 154. The central sequence control 156is further configured to control the controllable switches 68 a and 68b, to control the controllable switches 75 a and 75 b and hence thecontrollable resistive circuit 74, to control the controllable switches22 c and 22 d and hence the controllable resistive circuit 82 and tocontrol the controllable voltage source 84.

Here, the central sequence control 156 is configured to determine, basedon a current value analysis of a current time interval, for example anactual current rise velocity or an actual current drop velocity, setvalues for a next switching period and to determine, in the currentswitching period, the set values based on the actual values of theprevious switching period, respectively. A switching period can have,for example, a time period of less than 50 ns, less than 100 ns or lessthan 200 ns. In other words, the central sequence control 156 isconfigured to perform control of the output stages for the switch-onsequence, the switch-off sequence, the fixed and variable switch-onvoltage as well as the fixed switch-off voltage, the variable switch-offimpedance, the configuration of the di/dt-Monitor and the dynamic changeof the control times. The same is embodied, for example, as trailingdigital control loop. A digital control loop can have a directfeedthrough (durchgriff), such that a sequence or time portion canalready be adapted in the current period (k). Since it is hardlypossible, especially for fast switching processes in the 100 nsec range,to realize immediate control of the switching sequences, the approach oftracking control with direct feedthrough can be used here. This meansthat in the switching period k all control parameters are adaptedaccording to the switching behavior in switching period k−1 and are usedin the switching period k for the control. According to the controldeviation for di/dt as well as the timings for the control portions, thecontrol parameters are changed and used for the next switching periodk+1. The timings can be adapted based on the direct feedthrough alreadyfor the current switching period k. The control parameter adaptation isperformed independently both for the switch-on sequence and for theswitch-off sequence.

In other words, the VonMid-Control 58 represents the executing unit of aConfig VonMid-Modul 142 and readjusts the current change velocity in theswitch-on phase. Here, for example, the approach of adjustable sourcevoltage with fixed resistance Rgon is used. As is known, the currentchange velocity can be adjusted in the active region by the voltagechange velocity of the gate voltage. Thus, the same can be varied viathe change of the gate control current in this time period. Thus, withfixed resistance Rgon, the gate current can be influenced by changingthe driving source voltage. By the feedback of the di/dt-Monitor 94during the last switch-on edge, it has been determined whether thecurrent change velocity has been greater than the set value or smaller.If the set value becomes smaller, the control voltage for the di/dtportion is increased by at least one stage. Thus, in the next switch-onportion, the gate current rises and, hence the current change velocityrises.

However, if in the last switch-on edge a current change velocity thatwas greater than the set value has been determined, the control voltagefor the di/dt section is reduced by at least one stage. Thus, in thenext switch-on portion, the gate current is reduced and, hence thecurrent change velocity is reduced.

In one implementation option, control voltage adaptation is performed bymeans of a digitally adjustable voltage regulator. The output voltage ofthe same is, for example, recalculated and adapted in 256 stages by thedigital core, i.e., the control device 56 after passing each switch-onedge.

According to the switch-on portion, the Von-Control 66 sets thenecessitated driving voltage of the Gate-On paths. Up to the end of thedi/dt phase during the switch-on edge, Von-Control connects thedynamically adjustable control voltage VonMid to the control paths.Then, switching to the static on-voltage VCC is performed and the gateis operated with this voltage via the respective control paths.

Thus, it is also possible to set the control output stage in acompletely voltage-free state and, hence, prevent undesired switching-onof the power switch by errors in the downstream control paths.

According to the current switch-on portion, the Ron-Control 76 sets thematching resistance path configuration. In a first implementationoption, the same consists of two parallel switchable resistance pathshaving fixed values. Basically, four configuration options result. Forswitch-on portions I and IV, both paths are active simultaneously. Inportion II only path k1.1 and in portion III only path k1.2. Thereby,the di/dt portion can be controlled independently of the du/dt portion.In a further implementation option, the number of independent paths canbe extended as needed.

According to the current switch-off portion, the ROff-Control 82 setsthe matching resistance path configuration. In a first implementationoption, the same consists of two parallel switchable resistance pathshaving fixed values. Basically, this results in 4 configuration options.For switch-off portions I and IV, both paths are active simultaneously.In portion II, only path k2.1 and in portion III only path k2.2.Thereby, the di/dt portion can be controlled independently of the du/dtportion. In a further implementation option, the number of independentpaths can be extended as needed.

According to the switch-off portion, the VOff-Control 84 sets thenecessitated driving voltage of the Gate-Off paths. In a firstimplementation option, up to the end of the di/dt phase during theswitch-off edge, Voff-Control connects the voltage VDD as voltage sinkto the control paths.

In this option, V_(DD) is the supply voltage of the digital core as wellas LV periphery, i.e., the further active circuit parts of the device50. Thus, a large part of the charge fed into the gate of the powerswitch during switching-on is “recycled”. This represents a smart optionof minimizing the energy consumption of the GDU. Recycling can work whenthe voltage V_(DD) is smaller than the gate threshold voltage of thepower switch. Then, switching to the static OFF voltage V_(EE) isperformed and the gate is discharged completely up to the defined OFFvoltage V_(EE).

Thus, it is also possible to set the control output stage into acompletely voltage-free state and hence to prevent undesiredswitching-on of the power switch by errors in the downstream controlpaths.

The IOff-Control 88 represents the control instance for adjusting thecurrent drop velocity. In a first implementation option, for example,the approach of gradually adjustable impedance/current sink is used.

During switch-off portions I and II, the impedance is configured to theminimum value and merely the fixed resistor of the paths in theROff-Control is effective. When the di/dt portion is entered, theimpedance is configured to a value that leads to the desired currentdrop velocity.

It is known that the current drop velocity in the active area can beadjusted by the voltage change velocity of the gate voltage. Thus, thesame can be varied by varying the gate control current in this timeperiod. Thus, with fixed discharge voltage sink, the gate current can beinfluenced by changing the discharge resistance RGoff. By the feedbackof the di/dt-Monitor 94 during the last switch-off edge, it has beendetermined whether the current drop velocity was greater or smaller thanthe set value. If the actual value was smaller, the adjustable impedanceis reduced by at least one stage for the di/dt portion. Thereby, in thenext switch-off portion, the gate current is increased and the currentdrop velocity is increased.

If, however, in the last switch-off edge, a current change velocitygreater than the set value has been determined, the adjustable impedancefor the di/dt portion is increased by at least one stage. Thereby, inthe next switch-off portion, the gate current is reduced and hence thecurrent drop velocity is reduced.

In one implementation option, the impedance change is performed by meansof a digital-controlled transistor in analog operation. The forwardresistance of the same is recalculated and adapted, for example, in 256stages by the digital core after passing each switch-off edge. In analternative implementation option, the number of stages can be increasedfurther. In an alternative implementation option, the adjustable pathcan be a completely independent path without dynamical switching of theimpedance from switching operation to analog operation.

The Active GateClamp 102 is part of the security concept and representsa protective instance against Miller-induced switching-on of the powerswitch 12. In a first implementation option, the same is only effectiveduring the switch-off sequence and in the deactivated state. The same isrealized by a short-circuit transistor 104 connected between the gatecontrol line and the reference potential GND of the GDU. The same isactivated by an inadmissible voltage drop at a sensor resistor Rs. Ifthe potential on the side of the power switch is by at least a thresholdamount higher than the driver side potential, high-energy feedback, forexample by Miller coupling, exists. This can result in undesiredexceeding of the gate threshold of the power switch and the same becomesconductive again. The Active GateClamp prevents that by dynamicallyshort-circuiting the control gate.

The ActiveOff 108 is part of the security concept and represents afurther instance operating as static short-circuiter of the control gateterminal. However, the same does not act dynamically in an independentmanner like the Active GateClamp 102 but is controlled by the digitalcore. The gate voltage is detected by the Vge/Vgs-Monitor 122. If thegate voltage falls below a configured threshold, this is indicated tothe digital core and the ActiveOff Controller 144 activates theshort-circuiter Active GateClamp 102. A voltage level where theswitch-off process is terminated and merely the gate voltage is to bedischarged further is defined as a threshold in order to increase theinterference reserve. Thus, ActiveOff 108 supports the switch-offsequence, at the earliest starting from the switch-off portion IV andacts as discharge accelerator and at the same time as low impedancestatic short-circuiter.

The Active di/dt-Off-Clamp 112 is part of the security concept andrepresents a superordinate protective instance against inadmissibly highcurrent drop velocities. In the case of an error in the sequence controlof the digital core or error cases in the load circuit of the powerswitch 12 resulting in current drop velocities significantly above thedefined set value, the di/dt-Off-Clamp 112 becomes active and acts asinverse feedback branch in order to limit the current steepness.Contrary to normal control, this intervention is lossy.

The digital-controlled active gate drive unit (DCAGDU) can include thefollowing functional units: digital core, di/dt-Monitor, VonMid-Control,Von-Control, Ron-Control, Roff-Control, Voff-Control, Ioff-Control,Active GateClamp, ActiveOff, Active di/dt-Off-Clamp, Activedi/dt-On-Clamp, di/dt-Scaling, di/dt-Sens as well as Vge/Vgs-Monitor.

FIG. 6 shows a schematic illustration of the semiconductor-based switch12 where a first sensor 31 a for detecting the current change velocityof the power current flowing through the semiconductor-based switch 12is arranged at the first power terminal 36 a and wherein a second sensor31 b for detecting the current change velocity of the power current isarranged at the second power terminal 36 b. The first power terminal 36a, can, for example, be a source terminal of a MOSFET or a collectorterminal of an IGBT. The second power terminal can, for example, be adrain terminal of the MOSFET or an emitter terminal of the IGBT.Alternatively, the first power terminal 36 a and the second powerterminal 36 b can be exchanged. The sensors 31 a and 31 b, respectively,each include an insulating foil 32 a and 32 b, respectively as describedfor the sensor 31. On the insulating foil 32 a and 32 b, respectively,inductances 34 a and 34 b and 34 c and 34 d, respectively, are arrangedand are configured to detect the magnetic field at the respective powerterminal 36 a and 36 b, respectively. The insulating foils 32 a and 32b, respectively, include the mounting portion 45 a and 45 b,respectively, which is configured to allow an assembly at the respectivepower terminal 36 a and 36 b, respectively. For example, a current feedor a current drain, such as a cable, can be mounted, for example,screwed on the power terminal 36 a and 36 b, respectively, such that therespective assembly, such as the screw connection can be used for fixingthe sensor 31 a and 31 b, respectively at the power terminal 36 a and 36b, respectively.

A current direction of the power current can be, for example, arrangedperpendicular to a plane where the insulating foil 32 a and 32 b isarranged in space. A current detection direction of the inductances 34a-d can be arranged, for example, parallel to a viewing plane. Thismeans that the preferential direction 44 which is the same for allinductances 34 a-d in this implementation option, is perpendicular tothe current flow direction in space.

The sensors 31 a and 31 b, respectively, comprise connecting terminals,i.e. connecting points 166 a and 166 b and 166 c and 166 d,respectively, that are configured to provide the measurement voltage ofthe respective sensor 31 a and 31 b, respectively. The connectingterminals 166 a-d are arranged on the insulating foils 32 a and 32 b.This means that a signal provided by the inductances 34 a-d can beguided on or in the insulating foil 32 a or 32 b, for example, by meansof a conductive trace, to the respective connecting terminal 166 a-d andin that way a distance between a measurement line and the power terminal36 a and 36 b, respectively, can be increased. In that way, theconnecting terminals 166 a-d can be arranged spaced apart from theinductances 34 a and 34 b, respectively. This allows increasedelectrical insulation and reduction of spurious influences. For example,the connecting terminals 166 a and 166 b and 166 c and 166 d,respectively, can be connected to a shielded or non-shielded two-wireline that is configured to transmit the respective measurement voltage.The terminals can be mechanically and electrically fixed to the two-wireline, for example by means of a screw or solder connection.

Alternatively, the sensors 31 a and 31 b can also be configureddifferently. One of the sensors 31 a or 31 b, for example, can have oneof two different numbers of inductances 34 a/34 b and 34 c/34 d,respectively. Alternatively or additionally, one or several inductancescan have a preferential direction differing from the preferentialdirection 44.

In other words, the di/dt-Sens (sensors 31 a and 31 b, respectively)represents a central sensor instance which is both part of the normalcontrol and part of the security concept. In a first implementationoption, the sensor can be implemented as external measurement means inthe form of a small printed circuit board foil 32 a and 32 b,respectively. This foil 32 a and 32 b, respectively, fulfills therequirements of functionalization and can be mounted directly onto theterminal of the load emitter/load source of the power switch which isconnected, for example, at the intermediate circuit, for example, at theminus potential and can be screwed together with the terminal 36 a and36 b, respectively by means of a screw hole (mounting portion) 45 b_1and 45 b_2, respectively.

A di/dt sensor 31 a and 31 b, respectively, can also be arranged at thepower switch that is connected, for example, to the load collector/loaddrain at the intermediate circuit, for example, at the plus potential,and can be screwed onto this terminal 36 a and 36 b, respectively. Thus,each power circuit can comprise a di/dt sensor according to itsarrangement in the topology of the power part that is arranged in therespective commutation circuit. As an actual detection element, a woundinductance is used in the first implementation option. The size of theinductance is freely selectable and depends on the necessitated outputsignals. The inductance 34 a/34 b and 34 c/34 d, respectively, can bearranged as individual component or as a series connection. A memberhaving a ferrite core as a field concentrator is advantageous. Theinductance(s) 34 a/34 b and 34 c/34 d, respectively, can be implementedas a commercially available surface mounted device (SMD) ferritethrottle in SMD design or can also have any other design such as in athrough hole technology (THT). The inductances 34 a/34 b and 34 c/34 d,respectively, can have any inductance value, for example, 100 μH, 1 mHor 10 mH.

The distance of the sensor or the inductance to the current rail onwhich the sensor board is mounted as well as the orientation of theinductance with regard to the current direction can have a relevantinfluence. If possible, the orientation can be perpendicular to thecurrent direction in order to detect the maximum field. Wth respectivespacing sheets below the sensor foil, i.e. between the insulating foil32 a and 32 b, respectively, and the semiconductor-based switch 12, theinsulation ability can be increased and/or scaling of the signalamplitude can be performed, since the field strength and hence themeasurement voltage decreases with the distance of the sensor 31 a and31 b, respectively, to the current rail.

The sensitivity can hence be adjusted both via the selection parameterslength, width, number of turns, flow concentrator, number of members inseries of the inductance as well as by varying the distance betweensensor foil and current rail.

The di/dt sensor 31 a and/or 31 b can be connected to the GDU, i.e. thecontrol device, by a suitable two-wire connection via the connectingterminal 166 a/166 b and 166 c/166 d, respectively. This can be made,for example, as a pure two-wire connection. The two-wire connection canbe twisted in order to increase interference susceptibility.Alternatively or additionally, the connection can be made as shieldedtwo-wire connection with a shielding connection on a suitable potentialon the GDU.

For processing the measurement voltage further, the di/dt sensor 31 aand 31 b, respectively, is provided with high frequency (HF)/lowfrequency (LF) scaling (matching circuit) on the GDU (di/dt scaling).Thereby, both the signal termination of the measurement voltage as wellas the adaptation of the measurement amplitude to the further processinginstances can be enabled. The termination and the scaling, respectively,can be realized via a passive resistive and/or capacitive voltagedivider on the GDU. Thereby, a signal offset by an active signaladaptation can be prevented. Alternatively, scaling can be performed byan active impedance conversion. This enables the scaling and adaptation,respectively, of the same. Alternatively or additionally, scaling canalso be performed by means of a digitally adjustable transmission ratio,for example, by means of a programmable gate array (PGA) or by means ofa digital potentiometer.

Alternatively, the power terminal 36 a can also be arranged on a firstsemiconductor-based switch 12 and the power terminal 36 b on a secondsemiconductor-based switch 12, wherein the two semiconductor-basedswitches 12 are connected to a half bridge configuration. A load can bearranged between the two semiconductor-based switches.

It is an advantage of the above described aspects, both individually aswell as in a combination of several aspects, that a compact design ofcircuits can be obtained for cost-effective and simple integrationoption of the systems. Further, a number of active and passivecomponents regarding the obtained functionality is low which results inhigh reliability and high cost efficiency. Implementation andcombination, respectively, of the above-described aspects with adigitally configurable functional core allows the realization offunctions that have so far been difficult or not possible, as well aseasy reconfiguration, i.e., adaption to other applications, edgeparameters and/or functional characteristics and hence a high degree ofreusability and a high matching freedom, such that tedious and possiblyexpensive new development or redesign of a control can be prevented whena different type of switch is to be controlled. Wth the help of asensor, a measurement instance that is independent of the power switchmodule, i.e., the semiconductor-based switch, is provided, which is notbased on the parasitic inductances within the switch. A respective formfactor can be adapted to a high degree to a respective terminalconnection (connection of the semiconductor-based switch).

Omitting high-performance analog active components saves signal delaytimes, component costs and enables reduced energy consumption.

Status determination based on the sensor according to the third aspectenables prevention of using the collector or drain potential for statusand/or protective purposes. This enables prevention of using inaccurateand possibly a great number of high-volt (HV) components possibly havinghigh installation space requirements and additionally necessitate, dueto the necessitated voltage clearances, unused installation space. Insome HV switches, such a sufficient installation directly on therespective form factor of the GDU assembly according to conventionaltechnology is not possible, such that its secure operation is onlyfacilitated according to the above-described aspects.

The described readjusted control of the switching processes enablesoptimum behavior for almost any operating point and can hence save asignificant part of the switching losses. The described directfeedthrough allows adaptation of the switching behavior already duringthe current switching period. Switching, i.e., the switching behaviorcan be executed in a softer way at the portion boundaries, such thatinterference radiation during operation is reduced and, for example,standard requirements can be met or even higher standard requirementscan be met. This allows the admission of possibly expensive externalfilter measures.

One or several signal interfaces for superordinate control, such asRX-IF 124 and/or TX-IF 128 allow cross-wise watchdog functionality whichcan result in increased reliability. Implementing the second aspect inthe GDU enables internal consumption savings by recycling the controlenergy of the semiconductor-based switch. The possible reconfigurationof the switching parameters at runtime allows superimposed adaption ofthe behavior.

FIG. 7 shows a schematic block diagram of an electric circuit 70comprising a commutation circuit having a first voltage potential U₁,for example a plus potential, and a second voltage potential U₂, forexample a minus potential. This means that the electric voltage U isapplied between the potentials U₁ and U₂. The electric circuit 70includes two semiconductor-based switches 12 a and 12 b that areconnected to a half bridge circuit. The half bridge circuit comprises aload terminal between the semiconductor-based switches 12 a and 12 bthat is connected to a load impedance 172, for example of an electricconsumer. By means of the semiconductor-based switches 12 a and 12 b,the load impedance 172 can be switched between the potentials U₁ and U₂with regard to a potential U₃.

The load impedance 172 can be an impedance of any electric consumer, forexample an electric motor, an actuator, a sensor or, for example, alight or radiation source. Alternatively, the load impedance can alsothe be the impedance of a generator that can be switched between thepotentials U₁ and U₂ by means of the semiconductor-based switches 12 aand/or 12 b.

A device 10 a is configured to control the semiconductor-based switch 12a. Further, a device 10 b is configured to control thesemiconductor-based switch 12 b. The sensor 31 a is arranged at thepower terminal 36 a, such as a load collector or a load drain terminalof the semiconductor-based switch 12 a, wherein the power terminal 36 ais connected to the potential U₁. The sensor 31 b is arranged at thepower terminal 36 b, for example the load emitter or a load sourceterminal of the semiconductor-based switch 12 b, wherein the powerterminal 36 b is connected to the potential U₂.

Alternatively, instead of the device 10 a and/or 10 b, a device 20 canbe arranged to control the semiconductor-based switch 12 a and/or 12 b.Furthermore, merely one device 10, 20 or 50 can be arranged in order tocontrol both switches 12 a and 12 b. Alternatively, the load impedance172 can also be switched merely by means of switch 12 a or 12 b withregard to one of the potentials U₁ or U₂, i.e., alternative devicescomprise, for example, merely one semiconductor-based switch 12 a or 12b that is connected between the load and the potential U₁ or U₂.

FIG. 8 shows schematic diagrams of current and voltage curves,respectively, at the semiconductor-based switch during an activationprocess. The semiconductor-based switch is, for example, an IGBT. Forthe IGBT, a curve of the gate voltage (U_(Gate)), a curve of thecollector current I_(C) and a curve of a collector emitter voltageU_(ce) is plotted on a common time axis t. The time axis t has five timeinstants t₁, t₂, t₃, t₄ and t₅ that are arranged on the time axis t inthe stated order. The activation interval I (portion AI) is arrangedbetween time intervals t₁ and t₂. The activation potential AII isarranged between time intervals t₂ and t₃. The activation interval III(portion AIII) is arranged between time intervals t₃ and t₄. Theactivation interval IV (portion IV) is arranged between time intervalst₄ and t₅. This means that the activation intervals I to IV describedifferent portions (AI to AIV) of the activation process of thesemiconductor-based switch. In the different portions AI to AIV, thecontrol device can be configured to control the semiconductor-basedswitch differently based on different control targets. Depending on theactivation interval I to IV, the semiconductor-based switch can beinfluenced by different physical effects which can result in differentcontrol targets of the respective control devices, such as the controldevice 24 or the control device 56. Based on the control targets, therespective control device 24 or 56 can be configured to adjust a varyingcircuit configuration at an electric circuit or further components inorder to directly or indirectly influence the gate voltage.

The activation interval I (portion I) is represented by the delay timetd1,on. The voltage U_(ce) and the current I_(C) remain almost unamendedduring this time. This means that during a change (rise) of the gatevoltage U_(Gate) below the threshold voltage U_(g)(th), a conductivityof the switch remains almost unamended. If the semiconductor-basedswitch has the configuration “normally non-conducting”, the switch is inthe open state in this portion AI. The gate voltage UGate can bedetermined by the control device based on the measurement voltage of acurrent sensor, for example the current sensor 31. The determination canbe made directly, for example by a computing operation, indirectly, forexample in that the control device is configured to compare the currentchange velocity with a predefined threshold.

If the gate voltage U_(Gate) reaches the threshold voltage U_(g)(th),the semiconductor-based switch is configured to go into a conductivestate, this means the power current I_(C) begins to rise. Based oneffective leakage inductances in the semiconductor-based switch, thevoltage U_(ce) drops to a lesser degree. In portion AI, a control targetcan be described in a delay time that is as short as possible, thismeans it is a target to increase the gate voltage U_(Gate) in a timeperiod that is as short as possible. With reference to FIGS. 5a and 5b ,for example, the voltage of the controllable activation voltage sourcecan be in the range of the Miller voltage. A configuration of thecontrollable resistive circuit 76 set in the activation path 48 can bemade, for example such that both resistors k1,1 and k1,2 are effective,this means that the ohmic resistance Rgon is formed of a parallelconnection of the resistors k1,1 and k1,2. The transition into theportion AII (portion change trigger) is triggered when the gate voltageU_(Gate) is greater than the threshold voltage U_(g)(th) and/or a sensvoltage Usens (di/dt), for example o the sensor 31, is smaller than atrigger voltage Utrig,off, which can have a lower positive voltagevalue. The control device is, for example, configured to compare thetrigger voltage to the measurement voltage. Alternatively, thetransition to portion AII can also be triggered when the gate voltage isgreater than a trip threshold of the semiconductor-based switch 12.

In the activation interval II (portion AII), the current change velocitydi/dt is essential. The current I_(C) rises up to a current peak I_(C1),the voltage U_(ce) drops according to the parasitic inductances. Thismeans that the conductivity of the semiconductor-based switch increases.A control target of the control device in the portion AII can be, forexample, obtaining an edge of the voltage U_(ce), that is as flat aspossible, for reducing the return current peak as well as a lower fadingoscillation and a lower interference radiation of thesemiconductor-based switch. A respective circuit configuration can bemade by the control device such that the voltage of the controllableactivation voltage source is set in the range of the Miller voltage inorder to obtain a soft return current peak due to the dropping currentchange velocity.

During the portion AII, the respective controllable resistive circuit,for example the controllable resistive circuit 18 or 74 can beconfigured to effectively switch an ohmic resistance k1,1 or k1,2optimized, i.e. configured for the portion AII, for example the onehaving the greater resistance value. Transition into the followingactivation interval AIII can be described by a portion change triggerthat is triggered when the gate voltage U_(Gate) is greater than theMiller voltage U_(g)(mil) and/or when the sens voltage Usens (di/dt),such as the measurement voltage 42 is smaller than the trigger voltageUtrigg,on. The control device is configured to check whether a currentchange velocity (current rise velocity) occurs. Further, the controldevice is configured to compare a value of the current change velocityto the trigger voltage. Alternatively, the transition into the portionAIII can also be triggered when the gate voltage is smaller than a(small) trip threshold of the semiconductor-based switch 12.

The activation interval III (portion AIII) can be described such thatthe same determines the voltage drop velocity dU_(ce)/dt. The currentI_(C) can run according to the external load, for example the loadimpedance 172. The voltage U_(ce) can drop in the region of the staticminimum U_(ce,sat). It can be a control target of the respective controldevice to adjust an edge of the gate voltage U_(ce) that is as steep aspossible for reducing the switching losses, but at the same time toobtain a lower interference radiation. A respective circuitconfiguration can be, for example to adjust the voltage of thecontrollable activation voltage source, for example the controllablevoltage source 16 or 58 in the range of the voltage U_(g)(on), thestatic ON-voltage, in order to obtain a great dU_(ce)/dt. The voltageU_(g)(on) can have a value in a range whose bottom limit is greater thanthe Miller voltage U_(g)(mil) and whose upper limit is determined by amaximum gate voltage U_(g)(on)max of the semiconductor-based switch.Further, the control device can be configured to control thecontrollable resistive circuit such that an ohmic resistance optimizedfor the portion AIII is effective, for example the resistance k1,1 ork1,2 having the smaller resistance value. This can be performed, forexample due to a rising gate collector capacitance C.

A portion change trigger prior from the activation interval III to theactivation interval IV (portion AIV) can be triggered when the gatevoltage U_(Gate) is greater or by potencies greater than the Millervoltage U_(g)(mil) and/or a time interval Δtcon,mil, whose time periodcan be stored, for example in the timing register 132, has expired. Thismeans that runtime adaptation, for example with different intermediatecircuit voltages can be performed. In the activation interval IV(portion IV), the delay time td2,on is significant. During that time,the voltage U_(ce) drops to a static saturation value U_(ce,sat). Acontrol target of the control device can, for example be a delay timethat is as short as possible for reducing switching losses. A respectivecircuit configuration, can be made, for example such that the voltage ofthe controllable activation voltage source is set in the rangeU_(g)(on). The switch-on resistance Rgon can be performed by a parallelconnection of the two resistances k1,1 and k1,2, i.e. a parallelconnection of the resistances optimized for the portion AII (Ron di/dt)and AIII (Ron du/dt), respectively.

At the end of the activation interval IV, the semiconductor-based switchis in a static ON-state followed by a deactivation process (for examplea switch-off process). For example, during the static ON-state, it canbe determined via the sensor 31 by the control device whether ashort-circuit current flows through the semiconductor-based switch 12.Such an error case can be transmitted to a superordinate primary controlinstance by the control device. Alternatively or additionally, thecontrol device can be configured to initiate a switch-off ordeactivation sequence.

Alternatively, transition to portion AII, AIII or AIV can be performedwhen a respective time interval has expired, such as a stored timeperiod with regard to the portion AII (td1, on), a time period of theportion AII, Δtcon, mil or td2,on. If the control device determines thata respective control target of the portion is reached within a controlinterval, adaptation of the time interval, i.e. extending or reducingthe same, can be performed, for example by storing the current values inthe timing register. Further, based thereon, transition to thesubsequent portion can be performed.

In other words, the Digital Core (control device) represents theintelligent instance of the DCAGDU. The same can be implemented as acomplex programmable logic device (CPLD), FPGA, DSP or the like. Themain target of the driver (DCAGDU) is, for example the switch-on edgecontrol and switch-off edge control for each operating point that is asoptimum as possible in order to keep switching losses as low aspossible, to operate the power switch in the secure operating range, toload the complementary free-wheeling diode with as little reversecurrent as possible and to obtain an interference radiation behaviorconforming to standards. The DCAGDU can thus implement the approach ofportion-by-portion gate control with feedback of the switching statebased on the di/dt sensors and controlled tracking of the controlparameters.

The basic sequence of this driver unit divides the switch-on phase ofthe power switch in four portions: in portion A1, maximum driver poweris one target in order to minimize switch-on delay time. In portion A2,it is a target to obtain adapted driver power in order to remain in thedesired di/dt (current change velocity) for each operating point. Inportion AIII, for example, it is a target to obtain high defined driverpower in order to obtain the du/dt in the desired range andsimultaneously to pass through the Miller plateau as fast as possiblefor minimizing losses. In portion AIV it is a target to obtain maximumdriver power in order to pass through the Tail phase as fast as possibleand to obtain static saturation.

FIG. 9 shows the voltage curves U_(Gate) and U_(ce) as well as thecurrent curve for a deactivation process, for example a switch-offprocess that can be arranged prior to the activation process shown inFIG. 8 or, as an alternative, after the same. The time instants t₆, t₇,t₈ and t₁₀ are arranged on the time axis in the stated order. Thedeactivation interval DI is arranged between time intervals t₆ and t₇.The deactivation interval DII is arranged between time intervals t₇ andt₈. The deactivation interval DIII is arranged between time intervals t₈and t₉. The deactivation interval DIV is arranged between time intervalst₉ and t₁₀.

The deactivation interval DI (portion DI) represents the delay timetd,off. During this interval, voltage U_(ce) and current I_(C) remainessentially unamended. A possible control target of the respectivecontrol device is a delay time that is as short as possible. Arespective circuit configuration can, for example, be to adjust thevoltage of the controllable deactivation voltage source in the range ofthe Miller voltage. If the device comprises a controllable resistivecircuit in the deactivation path, such as the controllable resistivecircuit 82, the respective switch-off resistance Rgoff can be formed ofa parallel connection of the ohmic resistances k2,1 and k2,2, whereinone of the resistances k2,1 and k2,2 is optimized for one of theportions DII (Rgoff du/dt) or DIII (Rgoff di/dt). A portion changetrigger for transition into the deactivation interval DII can betriggered when the voltage U_(Gate) has approximately dropped into therange of the Miller voltage U_(g)(mil) and/or a time interval Δtd,offhas expired.

The deactivation interval II (portion DII) is determined by the voltagerise velocity dU_(ce)/dt. The voltage U_(ce) rises up to a value UZK(intermediate circuit voltage). A target of the control device 56 can beimplemented as a plateau retention time that is as short as possible(time in which the voltage Ugate is in the range of the Miller voltageU_(g)(mil)) and a relatively flat edge for reducing the interferenceradiation and the feedback effects. A respective circuit configurationcan be made, for example such that the voltage of the controllabledeactivation voltage source is set equal to a voltage V_(cc,off) and theeffective output resistance Rgoff du/dt is set and controlled,respectively, to the resistance k2,1 and k2,2 optimized for thisportion, for example the one having the smaller resistance value. Aportion change trigger for transition into the deactivation intervalDIII can be the fulfillment of the condition that the gate voltageU_(Gate) is smaller than the Miller voltage U_(g)(mil) and/or the sensvoltage Usens (di/dt) is smaller than a small, possibly negative,reference or trigger voltage Utrig,off1.

The trigger voltages Utrigg,on and/or Utrigg,off1 can have a low voltagevalue, which can have, for example 5%, 10% or 15% of the maximum gatevoltage. If the device is configured, for example, to bias the controlterminal in a negative manner, the trigger voltages Utrigg,on andUtrigg,off1 can also have a negative voltage value.

The deactivation interval 3 (portion DIII) determines the current dropvelocity dic/dt. The voltage U_(ce) rises according to the parasiticinductance of the semiconductor-based switch to a switch-off overvoltagevalue UOV. A target of the respective control device can be to obtain anedge of the voltage U_(Gate) and/or the current I_(c) that is as steepas possible. Further, a target can be to obtain low interferenceradiation as well as low switch-off overvoltage at the same time. Apossible circuit configuration for obtaining this target can beimplemented in that the voltage of the controllable deactivation voltagesource is set in the range U_(g)(th) or a voltage V_(cc,off) in order toobtain decreasing dic/dt. From the resistance effective at first duringportion DII, switching to the resistance optimized for the portion DIIIcan be performed at the beginning of the deactivation interval DIII,such as the one having the greater resistance value, i.e. for examplek2,2, such that the same is effective. A portion change trigger fortransition into the deactivation interval DIV can be triggered when thevoltage U_(g)(t) is smaller than the threshold voltage U_(g)(th) and/orthe sens voltage Usens (di/dt) is smaller than the positive triggervoltage +Utrig,off. For this, the control device can be configured tocheck whether a current change velocity (current drop velocity) occurs.Further, the control device is configured to compare a value of thecurrent change velocity to the trigger voltage or another referencevoltage.

The deactivation interval DIV (portion IV) determines the delay timetd2,off. During this time, the current I_(c) falls to its static value,for example 0 Ampere, 0.001 Ampere or 0.01 Ampere. One target of thecontrol device can be a delay time that is as short as possible forreducing the switching losses and for preventing renewed switching-on. Apossible switching configuration implementing that can be obtained whenthe voltage of the controllable deactivation voltage source is set to avalue V_(cc,off) and/or both switch-off resistances k2,1 and k2,2 areeffective, which means a parallel connection of the same is set in therespective controllable resistive circuit.

Alternatively, transition to portion DII, DIII or DIV can take placewhen a respective time interval has expired, for example a stored timeperiod with regard to td,off, a time period of the portion DII or DIII.If the control device determines that a respective control target of theportion is reached within a control interval, adaptation of the timeinterval can be performed, for example in that current values are storedin the timing register. Further, based thereon, transition to thesubsequent portion can take place. Alternatively, a time period of theportion D1, D2 and/or D3 can also be extended, for example when thecontrol target is not obtained, i.e. extension or reduction of therespective time period can be determined.

Compared to the methods of conventional technology, such a portionchange trigger allows intervention in the sequence control based ontriggered-based events. This means that contrary to purely time-basedcontrols, the state control of the semiconductor-based switch can beperformed based on detected states. This can be implemented as trackedcontrol with direct feedthrough for portion change corrections.

In other words, a target of the deactivation interval 1 (portion DI) isa maximum driver power in order to minimize the switch-off delay time.In a deactivation interval II (portion DID a target can be ahighly-defined driver power in order to keep du/dt in the desired rangeand at the same time pass through the Miller plateau as fast as possiblefor minimizing losses. In the deactivation interval III (portion DIII),an adapted driver power can be the target in order to remain within thedesired di/dt for each operating point. In the deactivation interval IV(portion DIV), a maximum driver power can be a target in order to passthrough the tail phase as fast as possible to reach the secure OFFstate. This means that also the switch-off phase can be divided intofour portions.

Recycling the charge carriers received from the control terminalaccording to the second aspect can be performed, for example, up to theend of the third portion DIII. Alternatively, switching to the staticOFF-voltage can also be performed at another time, for example duringthe third portion DIII. This can result in reaching the staticOFF-control voltage faster by reducing a concentration of the recycledcharge carriers. In other words, a change between source voltage VCC andstatic OFF-voltage VEE can result, during the portion DIII, in anaccelerated fading-off or the control limit for the maximum currentsteepness can basically be increased. A change at the beginning orwithin the portion DIV results in softer fading-off and an increasedportion of restored energy.

The above-described devices and/or control methods are suitable forregulated control of MOSFETs and IGBTs in hard-switching applications in2-level or multi-level-topologies. Basically, all possible modulepackages as well as customized and/or application-specific structureversions can be used. All voltage classes of a few 100 volts up tohigh-volt (HV) applications, for example in a range of more than 3.3 kV,more than 4.5 kV or more can be addressed. The solution is also suitablefor fast switching applications.

The presented implementation options describe a functional unit of adigital control core, which optimally controls, with the help of a di/dtsensor as feedback path, the switch-on and switch-off edge of a powerswitch for each operating point at runtime. This can be performed, forexample, with at least one resistive switch-on path combined with atleast one adjustable voltage source as well as with at least oneresistive switch-off path in combination with at least one adjustablevoltage source/sink. In embodiments, the information from the di/dtsensor serves to adapt the runtimes, control of the switching paths andregulation of the adjustable voltage sources/sinks. Further, embodimentsprovide simultaneous control option of the switch-off edge and feedbackof the control energy when switching off the semiconductor-based switchin a generally used auxiliary energy source of the GDU. This can berealized with at least one adjustable resistive switch-off path incombination with at least one stationary auxiliary energy source.

Further, the above embodiments according to the third aspect describe arealization of the di/dt sensor. The same is embodied in a specificimplementation option. Further, a dynamic gate voltage inverse feedbackis described as protection against undesired (renewed) switching-on dueto Miller-induced feedback of voltage changes at the power switch. Sucha realization can be provided by detecting the voltage differencebetween set value of the gate voltage and actual value of the gatevoltage at the power switch and can be provided to an active switch asinverse feedback path for reducing the actual value.

Although some aspects have been described in the context of anapparatus, it is clear that these aspects also represent a descriptionof the corresponding method, such that a block or device of an apparatusalso corresponds to a respective method step or a feature of a methodstep. Analogously, aspects described in the context of a method stepalso represent a description of a corresponding block or item or featureof a corresponding apparatus.

Depending on certain implementation requirements, embodiments of theinvention can be implemented in hardware or in software. Theimplementation can be performed using a digital storage medium, forexample a floppy disk, a DVD, a Blu-Ray disc, a CD, an ROM, a PROM, anEPROM, an EEPROM or a FLASH memory, a hard drive or another magnetic oroptical memory having electronically readable control signals storedthereon, which cooperate or are capable of cooperating with aprogrammable computer system such that the respective method isperformed. Therefore, the digital storage medium may be computerreadable. Some embodiments according to the invention include a datacarrier comprising electronically readable control signals, which arecapable of cooperating with a programmable computer system, such thatone of the methods described herein is performed.

Generally, embodiments of the present invention can be implemented as acomputer program product with a program code, the program code beingoperative for performing one of the methods when the computer programproduct runs on a computer. The program code may for example be storedon a machine readable carrier.

Other embodiments comprise the computer program for performing one ofthe methods described herein, wherein the computer program is stored ona machine readable carrier.

In other words, an embodiment of the inventive method is, therefore, acomputer program comprising a program code for performing one of themethods described herein, when the computer program runs on a computer.A further embodiment of the inventive methods is, therefore, a datacarrier (or a digital storage medium or a computer-readable medium)comprising, recorded thereon, the computer program for performing one ofthe methods described herein.

A further embodiment of the inventive method is, therefore, a datastream or a sequence of signals representing the computer program forperforming one of the methods described herein. The data stream or thesequence of signals may for example be configured to be transferred viaa data communication connection, for example via the Internet.

A further embodiment comprises a processing means, for example acomputer, or a programmable logic device, configured to or adapted toperform one of the methods described herein.

A further embodiment comprises a computer having installed thereon thecomputer program for performing one of the methods described herein.

In some embodiments, a programmable logic device (for example a fieldprogrammable gate array, FPGA) may be used to perform some or all of thefunctionalities of the methods described herein. In some embodiments, afield programmable gate array may cooperate with a microprocessor inorder to perform one of the methods described herein. Generally, themethods are performed by any hardware device. This can be a universallyapplicable hardware, such as a computer processor (CPU) or hardwarespecific for the method, such as ASIC.

While this invention has been described in terms of several advantageousembodiments, there are alterations, permutations, and equivalents whichfall within the scope of this invention. It should also be noted thatthere are many alternative ways of implementing the methods andcompositions of the present invention. It is therefore intended that thefollowing appended claims be interpreted as including all suchalterations, permutations, and equivalents as fall within the truespirit and scope of the present invention.

1. Electric circuit, comprising: a semiconductor-based switch with afirst terminal and a second terminal for conducting a power current; asensor for detecting a current change velocity of the power currentflowing through the semiconductor-based switch, the sensor comprising:an insulating foil configured to be connected to the first or secondterminal of the semiconductor-based switch; and an inductance arrangedon the insulating foil on a side of the same that is arranged oppositeto a side facing the semiconductor-based switch during a measurementoperation of the sensor in order to detect a magnetic field generated bythe power current and to provide a measurement voltage based on thedetected magnetic field; wherein the inductance is spaced apart from thesemiconductor-based switch during the measurement operation at least bythe insulating foil, such that contactless measurement of the currentchange velocity by the sensor is enabled; wherein the insulating foilfurther comprises a mounting portion; wherein the first or secondterminal of the semiconductor-based switch is connected to the mountingportion of the insulating foil; wherein the first or second terminal ofthe semiconductor-based switch is connected to a current conductor,wherein the mounting portion at least partly surrounds the currentconductor; and wherein the first or second terminal is connected to thecurrent conductor by means of a screw connection, wherein the mountingportion of the sensor is a screw hole, and wherein the sensor isarranged between a first and a second element of the screw connectionand is fixed by means of the screw connection.
 2. Electric circuitaccording to claim 1, wherein the inductance comprises at least onepreferential direction where a sensitivity of the inductance withrespect to the detected magnetic field is at a maximum, wherein theinductance is arranged on the insulating foil with an orientation withrespect to the preferential direction, wherein the preferentialdirection is arranged, within a tolerance range, perpendicularly to adirection of the power current when the same flows.
 3. Electric circuitaccording to one of the preceding claims, wherein the mounting portioncomprises a recess of the insulating foil, wherein the insulating foilat least partly surrounds the recess.
 4. Electric circuit according toclaim 1, wherein the insulating foil of the sensor comprises at leastone connecting terminal that is arranged spaced apart from the mountingportion and spaced apart from the inductance, wherein the sensor isconfigured to provide the measurement voltage at the at least oneconnecting terminal and wherein the sensor comprises a measurement linewhich is mechanically and electrically fixed to the connecting terminal.5. Electric circuit according to claim 1 comprising at least one Zenerdiode connected to a controllable switch and configured to switch thecontrollable switch into a closed state when the Zener diode isconductive, wherein the controllable switch is configured to connect, inthe closed state, a control terminal of the semiconductor-based switchto an inverse feedback potential, wherein the Zener diode is connectedto the sensor, such that the measurement voltage of the sensor can beapplied to the Zener diode, wherein the Zener diode comprises abreakdown voltage essentially corresponding to a threshold, such thatthe inverse feedback potential is effective at the control terminal whenthe measurement voltage exceeds or falls below the threshold. 6.Electric circuit according to claim 1 comprising a control device thatis configured to control the semiconductor-based switch based on themeasurement voltage.
 7. Electric circuit according to claim 6, wherein amatching circuit is arranged between the control device and the sensor,which is configured to change an amplitude of the measurement voltage.8. Device for switching a semiconductor-based switch, comprising: anelectric circuit according to claim 1; a controllable voltage sourcethat is configured to provide a time-varying voltage potential; acontrollable resistive circuit comprising at least two ohmic resistancesconnected in parallel that are controllable such that at least threeresistance values of the parallel connection result; a control devicethat is configured to control the controllable voltage source and thecontrollable resistive circuit independent of one another based on themeasurement voltage.
 9. Device according to claim 8, wherein thetime-varying voltage potential is galvanically coupled to a supplypotential of the control device and has a lower potential value than athreshold voltage of the semiconductor-based switch, wherein the controldevice is configured to control, during a passivation process duringwhich a state of the semiconductor-based switch is changed based on areduced concentration of charge carriers that are stored in a controlcapacitance of the semiconductor-based switch, the controllable voltagesource such that, based on the provided time-varying voltage potential,charge carriers that are stored in the control capacitance flow out ofthe control capacitance and contribute to an operation of the controldevice based on the galvanic coupling.
 10. Device according to claim 8,wherein the control device is further configured to control, in atime-varying manner, based on the measurement voltage, a furthercontrollable resistive circuit comprising at least two ohmic resistancesconnected in parallel that can be controlled such that at least threeresistance values of the parallel connection result, and a furthercontrollable voltage source, in order to control, based on themeasurement voltage, during an activation time interval in which aconcentration of charge carriers in a control capacitance of thesemiconductor-based switch is increased, an increase of theconcentration of charge carriers and wherein the control device isconfigured to control, based on the measurement voltage, during apassivation time interval in which the concentration of charge carriersis reduced, a drop of the concentration.
 11. Device according to claim8, wherein the control device is configured to compare, in a firstcontrol interval, the measurement voltage to a set value for acquiring acomparative result, and to control, in a second control intervalfollowing the first control interval, the controllable voltage sourceand the controllable resistive circuit based on the comparative result.12. Device according to claim 11, wherein the control device isconfigured to determine, during the control interval, information withrespect to a time period of a current control interval based on thecomparative result and to carry out a change between portions based onthe comparative result.